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LM1897N
18 V, 6 mA, low noise preamplifier for tape playback system
National .
Semiconductor
LM1897 Low Noise Preamplifi
for Tape Playback Systems
General Descri tion " Low Voltage Battery Operation 4V
Th M1897i IN E . lifi f li . a Wide gain bandwidth due to broadband
6 .L. .'s a dual 19 gain preamp: , r or IPP Icathns two amplifier approach 76 dB @ 20 kHz
requiring optimum nelse performance. It IS an ideal choice a Hi h ower su I reie cti on 105 dB
for a tape playback amplifier when a combination of low lg '.'. . ppy I tt
noise, high gain, good power supply rejection, and no power a Low distortion th03 Yo
up transients are desired. The application also provides a Fast slew rate SWPS
transient-free muting with a single pole grounding switch. ll Short circuit protection
a Internal diodes for diode switching applications
Features a Low cost external parts
a Programmable turn-on delay I: Excellent low frequency response
I: Transient-free power up-no pops n Prevents "click" from being recorded onto the tape
a Transient-free muting duriy ppwer supply cycling in tape playback
a Low noisty-0.6 p.V CCIR/ARM in a DIN circuit refer- applications
enced to gain at 1 kHz
Iftt ttht RI m " 1.5719 "
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In lot ins M CI 0.0022;§ "
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0,t tf C2 + - M m " H
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" " tt tt 10
‘on 15 N
1 x25 - - D", mm
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INPUI I0
INPUT + tot
xts T - --trl-r WM
4 , S 5 7
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FIGURE 1. Typical Tape Playback Preamplifier Application
Order Number LM1897N
See NS Package Number N165
TL/H/7094-1
£68lW'l
LM1897
Absolute Maximum Ratings
If MllltaryfAeroapatttt spttelfled devices are required, Storage Temperature -65'C to + 150°C
please contact the National Sqtrttlttondutttor Sales Operating Temperature 0°C to + 70°C
oftltatfDltttrlbutora for avaliablllty and speclflcatlons. Minimum V olta g 9 On Any Pin _ th1 VDC
Supply Voltage 18V Lead Temperature (soldering, 10 sec.) 260'C
Voltage on Pins 8. and 9 18V
Package Dissipation (Note 1) 715 mW
Electrical Characteristics (TA = 25°C, vcc = 121/, See Circuit-Figures,
Parameter Conditions Min Typ Max Units
Operating Supply Voltage Range Rs removed from circuit 4 18 V
Supply Current VCC = 12V 6 12 mA
Total Harmonic Distortion f = 1 kHz, VIN = 0.3 mV, Pins 7 & IO, Figurs2 0.03 M,
THD + Noise (Note 2) f = 1 kHz, VOUT = IV, Pins 7 &10, Figure? 0.10 0.25 %
Power Supply Rejection Input Ref. f = 1 kHz, 1 VRMS 85 105 dB
Channel Separation f = 1 kHz, Output = 1 VRMS, Output to Output 40 60 dB
Signal to Noise (Note 3) Unweighted 32 Hz-12.74 kHz (Note 2) 58 dB
CClR/ARM (Note 4) 62 dB
A Weighted 64 dB
CCIR, Peak (Note 5) 52 dB
Noise Output Voltage CClR/ARM (Note 4) 120 200 p.V
Input Amplifiers
Input Bias Current 0.5 2.0 WA
Input Impedance f = 1 kHz 50 kn
A.C. Gain 27 28 29 dB
A.C. Gain Imbalance 10.15 * 0.5 dB
D.C. Output Voltage 1.8 2.2 2.6 V
D.C. Output Voltage Mismatch Pins 3 and 14 - 200 i 30 + 200 mV
Output Source Current Pins 3 and 14 2 10 mA
Output Sink Current Pins 3 and 14 300 600 “A
Output Amplifiers
Closed Loop Gain Stable Operation 5 WV
Open Loop Voltage Gain D.C. 110 dB
Gain Bandwidth Product MHz
Slew Rate 6 V/ps
Input Offset Voltage 2 5 mV
Input Offset Current 20 100 nA
Input Bias Current 250 500 nA
Output Source Current Pin 7 or 10 2 10 mA
Output Sink Current Pin 7 or 10 400 900 “A
Output Voltage Swing Pin 7 or 10 11 Vpp
Output Diode Leakage Voltage on Pins 8 and 9 = 18V 0 10 MA
Note 1: For operation in ambient temperatures above 25'0. the device must be derated based on a 150'C maximum Junction temperature and a thermal resistance
ot 175'C/Wan junction to ambient.
Note Y. Measured with an average responding voltmeter using the filter circuit in Figure 4. This simple filttyr is approximately equivalent to a “brick wall" filter with a
passband of 20 Hz to 20 kHz (see "Application Hints" section). For 1 kHz THD the 400 Hz high pass rdter on the distortion analyzer is used.
Note & The numbers are referred to an output level of 160 mV at Pins 7 and 10 using the circuit of Figursz This txtrrtrsponds to an input level of 0.3 mV RMS at
333 Hz.
Note 4: Measured with an average responding voltmeter using the Dolby lab‘s standard CCIR fitter having a unity gain retevence at 2 kHz.
Note 5: Measured using the Rhode-Schwarz psophometer. model UPGR.
1lgg M INK
- " ttlt "
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th1 w Rt ttlk ' Fu-
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"''gllf'" l Illt tA tttt 1%
m- 1 V" "J. .
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ttlt m T M." Mttttsf "
- . 1%
I 'ttlk
FIGURE 2. General Test Clrcult
Gain vs. Frequency
30 ' , T _ '
CIRCUIT 0F FIGURE ,
a so 's,
g 50 k
til 50100200 6001K n " Wka
Fasaumcv ON
TL/H/7DB4-a
FIGURE 3. Froquoncy Response of Tom Circuit
INPUT o--
FIDM L311“?
PINS 1 tht "
'o'illi;','-,, ---t-,CCE
mm DIITDHTION ANALYZEI/
VOLTMETEI
FIGURE 4. Simple 32 Hz-12740 Ha Filter and Motor
TL/H/7094-2
TL/H/7094-4
LM1897
REGULATOR
External
Compo-
Component (Refer to
Figure n External
Component Function
Set turn-on delay and second
amplifier's low frequency pole.
Leakage current in C2 results in
DC offset between the amplifier’s
inputs and therefore this current
should be kept low. R1 is set equal
to H2 such that any input offset
voltage due to bias current is
effectively cancelled. An input
offset voltage is generated by the
input offset current multiplied by
the value of these resistors.
Set the DC and iow frequency gain
of the output amplifier. The total
input offset voltage will also be
multiplied by the DC gain of this
amplifier. It is therefore essential
to keep the input offset voltage
specification in mind when
employing high DC gain in the
output amplifier; Le. 5 mV X 400
= 2V offset at the output.
Set tape playback equalization
characteristics in conjunction with
R3 (calculations for the
component values are included in
the Applications Hints section).
TL/Hf7094-5
FIGURE 5. Schematic Diagram
Normal
of Value
2 kn -40 kn
0.1 pF-
Leakage)
40 kft
500 kn-
10 Mt-
200 kn
0.00047 PU'-
0.01 “F
External Component (Refer to Normal
Compo- Figure nExternal Range
nent Component Function of Value
H5 Biases the output diode when it is 2 kft-
used in DC switching applications. 47 kn
This resistor can be excluded if
diode switching is not desired.
Cs Often used to resonate with tape 100 pF-
head in order to compensate tor 1000 pF
tape playback losses including
tape head gap and eddy current.
For a typical cassette tape head,
the resonant frequency selected is
usually between 13 and 17 kHz.
Rs Increases the output DC bias 100 kit-
voltage from the nominal 2.2V 10 Mn
value (See the Application Hints
section).
R7 Optionally used for taps muting.
The use of this resistor can also
provide "No Pop" turn-off if
desired.
Application Hints
DISTORTION MEASUREMENT METHOD
In order to clearly interpret and compare specifications and
measurements for low noise preamplifiers, it is necessary to
understand several basic concepts of noise. An obvious ex-
ample is the measurement of total harmonic distortion at
very low input signal levels. Distortion analyzers provide out-
puts which allow viewing of the distortion products on an
oscilloscope. The oscilloscope often reveals that the "dis-
tortion" being measured contains 1) distortion, 2) noise, and
3) 50 or 60 cycle AC line hum.
Application Hints (Continued)
Line hum can be detected by using the "line sync" on the
oscilloscope (horizontal sync selector). The triggering of a
constant wave form indicates that AC line pickup is present.
This is usually the result of electro-magnetic coupling into
the preamplifier's input or improper test equipment ground-
ing, which simply must be eliminated before making further
measurements!
Input coupling problems can usually be corrected by any
one of the following solutions: 1) shielding the source of the
magnetic field (using mu metal or steel). 2) magnetically
shielding the preamplirer, 3) physically moving the pream-
plifier far enough away from the magnetic field, or 4) using a
high pass filter (to == 200 Hz-l kHz) at the output of the
preamplifier to prevent any line signal from entering the dis-
tortion analyzer. Ground loop problems can be solved by
rearranging ground connections of the circuit and test
equipment.
Separating noise from distortion products is necessary
when it is desired to find the actual distortion and not the
signal-to-noise ratio of an amplifier. The distortion produced
by the LM1897 is predominately a second harmonic. It is for
this reason that the third and higher order harmonics can be
filtered without resulting in any appreciable error in the mea-
surement. The filter also reduces the amount of noise in the
measured data. Another more tedious technique for mea-
swing THD is to use a wave analyzer. Each harmonic is
measured and then summed in an RMS calculation. A typi-
cal curve is plotted for distortion vs. frequency using this
method. A typical curve is also included using a 20 Hz to
20 kHz 4th order filter.
To specify the distortion of the LM1897 accurately and also
not require unusual or tedious measurements the following
method is used. The output level is set to one volt RMS at 1
kHz (approximately 5 millivolts at the input). The output is
filtered with the circuit of Figure 4 to limit the bandwidth of
the noise and measured with a standard distortion analyzer.
The analyzer has a filter that is switched in to remove line
hum and ground loop pick-up as well as unrelated low fre-
quency noise. The resulting measurement is fast and accu-
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques. they appear equal in measure-
ments. This is often the case when comparing integrated
circuit to discrete preamplifier designs. Discrete transistor
preamps often "run out of gain" at high frequencies and
therefore have small bandwidths to noise as indicated be-
DISCRETE
mm (an)
20 Nil 2k
28k 200k 2M
FREOUENCY
TL/H/7094-6
FIGURE 6
Integrated circuits have additional open loop gain allowing
aditional feedback loop gain in order to lower harmonic dis-
tortion and improve frequency response. it is this additional
bandwidth that can lead to erroneous signal to noise mea-
surements if not considered during the measurement pro-
cess. In the typical example above, the difference in band-
width appears small on a log scale but the factor of 10 in
bandwidth, (200 kHz to 2 MHz) can result in a 10 dB theo-
retical difference in the signal-to-noise ratio (white noise is
proportional to the square root of the bandwidth in a sys-
tern).
In comparing audio amplifiers it is necessary to measure the
magnitude of noise in the audible bandwidth by using a
"weighting" filter.1 A "weighting" filter alters the frequency
response in order to compensate for the average human
ear's sensitivity to certain undesirable frequency spectra.
The weighting filters at the same time provide the bandwidth
limiting as discussed in the previous paragraph.
The 32 Hz to 12740 Hz filter shown in Figure 4 is a simple
two pole, one zero filter, approximately equivalent to a
"brick wall" filter of 20 Hz to 20 kHz. This approximation is
absolutely valid if the noise has a flat energy spectrum over
the frequencies involved. In other words a measurement of
a noise source with constant spectral density through either
of the two filters would result in the same reading. The out-
put frequency response of the two filters is shown is Figure
P--""'""'"--"
NOISE A1
AMPLITUDE tun)
N FREDU ENCV
“BRICKWALL” FILTER
BAN DWIDTH
AMPLITUDE (dB)
N 32 12740 20k
32-12140 m FILTER
FIGURE 7
Typical signal-to-noise figures are listed for several weight-
ing filters which are commonly used in the measurement of
noise. The shape of all weighting filters is similar, with the
peak of the curve usually occurring in the 3 kHz-7 kHz re-
gion as shown below.
TL/ H/ NM- 7
AMPLITUDE
til too " 6k 20k
FREQUENCY
FIGURE 8
TL/H/7094-8
£68|~W1
LM1897
Application Hints (Continued)
In addition to noise filtering, differing meter types give differ-
ent noise readings. Meter responses include: 1) RMS read.
ing, 2) average responding, 3) peak reading. and 4) quasi
peak reading. Although theoretical noise analysis is derived
using true RMS (root mean square) based calculations,
most actual measurement is taken with ARM (Average Re-
sponding Meter) test equipment. Unless otherwise noted an
average responding meter is used for all AC measurements
in this data sheet.
BASIC CIRCUIT APPROACH
The LM1897 IC incorporates a two stage broadband design
which minimizes noise, attains overall DC stability and pre-
vents audible transients during turn-on.
The first stage is a direct coupled amplifier with an internal
gain of 25 V/V (28 dB). Direct coupling to the tape head
reduces input source impedance and external component
cost by removing the input coupling capacitor. A typical in-
put coupling capacitor of 1 psf' has a reactance of 1.5 kn at
100 Hz. The resulting noise due to the amplifier's input
noise current can dominate the noise voltage at the output
of the playback system. The input of the amplifier is biased
from a reference voltage that is temperature compensated
to produce a quiescent DC voltage of 2.2V at the output of
the first stage. The input stage bias current that flows
through the tape head is kept below 2 FA in order to pre-
vent any erasure of tape moving past the head. An added
advantage of DC biasing is the prevention of large current
transients during the charging of coupling capacitors at turn-
on and turn-oft.
The second stage provides additional gain and proper
equalization while preventing audible turn-on transients or
"pops". The output (Pin 10) is kept low until C2 charges
through RI. When the voltage on C2 gets close to the DC
voltage on Pin 14, the output rises exponentially to its final
DC value. The result is a transient-fret, turn-on characteris-
Internal diodes are provided to facilitate electronic diode
switching popular in automotive applications.
The general test circuit illustrates the topography of the sys-
tem. The components determining the overall frequency re-
sponse are external due to the extreme sensitivity when
matching tt DIN equalization curve.
MUTE CIRCUIT
The LM1897 can be muted with the addition of two resistors
and a grounding switch, as shown in Figure 1. When the
circuit is not muted the additional resistors have no effect on
the AC performance. They do have an effect on the DC C)
point however,
The difference in the DC output voltages of the input amplifi-
ers is applied across the mute resistors (R7) and the posi-
tive input resistors (R1). This results in an additional offset
at the input of the output amplifiers. To keep this offset to a
minimum R7 should be as large as possible to achieve ef-
fective muting. In all cases R7 should be at least ten times
RI. A typical value of R7 is 25 to 50 times R1.
CAPACITOR-COUPLED INPUT
The LM1897 is intended to be coupled directly to the signal
source. Direct coupling permits faster turn-on and less low-
frequency noise than would be possible with a capacitor-
coupled input. However, there are some applications which
require that the signal source be referred to ground and
coupled to the input through a capacitor, Figure 9 is an ex-
ample of an LM1897 with a capacitor-coupled input. As
shown, the circuit has a flat frequency response and is suit-
able for use as a microphone preamp.
Fig provides a DC path for input bias current. The value of
He should be as low as possible without loading the source.
A very large value of 853 can cause excessive DC offset at
the amplifier output. In order to avoid turn-on pops, the in-
verting input of the second amplifier must be at a higher
voltage than the non-inverting input when Voc is applied.
R10, RIS, R12, and D1 ensure that this condition will be met.
If later stages in the playback system employ turn-on muting
circuitry, these extra components may not be needed. The
value of R10 depends on Vcc as defined by the following
relationship:
R10 = (Vcc - I) X 1k
R12 R3
$6.5: -
TL/H/7094-9
FIGURE 9. Mlcrophone Preamplifier with Capacltor Coupled Input
Application Hints (Continued)
Design Equation
The overall gain of the circuit is given by:
-R4R3 ( 1 ) ( 1 )
A ---2sL-gy-ta, tt+_-, s+t---- 1
V Raina + H4) R461 (Ra + R4)C1 ( )
Standard cassette tapes require equalization of 3180 its (50
Hz) and 120 ps (1 .3kHz). These time constants result in an
AC gain at 1 kHz given by:
-R R 3180psor50Hz
Av(1 kHz) = 25 (s-re'-?,,-,)) . 3 and
at a + 4) 120psor1326Hz
Using the pole and zero locations of the transfer function,
the two other equations needed to solve for the component
values are:
R = -.-...- a
4 211010326 Hz) ( )
= _ = 4
Ra 2erth(50 Hz) 21rC1(1326 Hz) 27rC1(51.96) ( )
We can now solve for ch as a function of R2, or:
[2,rc,1-032, l [2,,d-sls, ]
Av(1kH2) = -25 217C1(1326) 1211C1(51.96) (1.663)
R ....._
[ 22nc1isoi] (5)
-4.80 x 10-3
= R2 [AV(1 kHz)l
When chromium dioxide tape is used, the defined time con-
stants are 3180 ps and 70 ps. This changes equation (3) to:
= -...--....._...-- 7
thrth (2274 Hz) ( )
The value of R3 is normally not changed. This results in an
error of less than 0.2 dB in the low frequency response.
The output voltage of the LM1897 is set by the input amplifi-
er DC voltage at pin 3 or 14, and by R3 and Rs.
Nominal VOUT (pin 7 or 10) = 2.2 (1 + %) (8)
Pins 8 and 9 are biased 0.7 volts less than VOUT (pin 7 or
10). When these diodes are used the output (pin 7 or IO)
should be biased at one half the minimum operating supply
voltage. Equation (8) can be rewritten to solve for Rs,
Vr,--2.t? (9)
The output voltage of the LM1897 will vary from that given
in equation (8) due to variations in the input amplifier DC
voltage as well as the output amplifier input bias current,
input offset current and input offset voltage. The following
equation gives the worst case variation in the output volt-
AVOUT = i [AVPIN 3 (1 + 'le) + ',-li-(sAsim- R2)) +
"373191 + Ra) + Vos)] (10)
Using the worst case values in the electrical characteristics
reduces this to
AVOUT= t [0.4(1 +1Fh) +
?(aoown, - Ra) + 50nA(R1+ Ra) + 5mV))] (11)
The turn-on delay is set by R, and Ca; delay can be approxi-
mated by:
DelayTimet = F11C2 In ( ) (J) (12)
Vooc R2
Example
If we desire a tape preamp with 100 mV output signal from a
tape head with a nominal output of 0.5 mV at 1 kHz for
standard ferric cassette tape, the external components are
determined as follows. The value of R2 is arbitrarily set to 10
R1 = R2 = 10k
This minimizes errors due to the output amplifier bias cur-
rents.
-4.80 x 10-3
- 100 mV]
0.5 mV
Use 0.0022 p.F and determine:
- 2nc1(1326)
- 2arC1(51.96)
To bias the output amplifier output voltage at 6 volts (half
supply):
th = = 2400 pF - 0.0022 11F
R4 = 54.6 kft - 54.9 k0. 1%
Rs ---1.39Mn-r1.4Mn1%
- 2.2(1.4 Mn)
6 - 2.2
The maximum variation in the output voltage is found using
equation (11):
Rs = 811 ktt-r820 kn
AVOUT = 11.9V
The low frequency response and turn-on delay determine
the value of Ca. For R1 == 10k and Ca = 10p.F the low
frequency 3 dB point is 1.6 Hz and the turn-on delay is 0.4
seconds. from equation (12).
The complete circuit is shown in Figure 2. A circuit with 5%
components and biased for a minimum supply of 10 volts is
shown in Figure 1. If additional gain is needed R1 and R2
can be reduced without changing the frequency response of
the circuit.
Reference 1: CClR/ARM: A Practical Noise Measurement
Method; by Ray Dolby, David Robinson and Kenneth Gun-
dry, AES Preprint No. 1353 (F-3).
£68 l “'1
LM1897
Typical Performance Characteristics
Total Harmonlc Distortion vs
Frequency PSRR " Vcc
EMU" ttF FIGURE ,
b' " Vom=1ms m
E s E m
i .2 t? 30
a A mm menu:
, w cmcun or nsua: t
. 60 mm“ -
g .02 'c--''ifhoum
.II M I I 1
ttt '0t0"tr0Mote" snout“ 4 6 8 " 12 " IS "
nmsucv m (vans)
Ict: " Supply Voltage PSRR vs Frequency
cannon or mun: 2
or neuan
Vcc=12V
' I I 10 " " " " " snmzoosoou " "tNat
ll“ tttt
SUPPLY VOLTAGE (VOLT!) Fft00EtttN tht)
Input Amplifier THD vs Input Amplifier Galn and
Input Level Phase vs Frequency
; PHASE ts
- ht = m so u so
t 1:1th "
g tit us so
tl a "
g E um so
i it 105
tt N " 50 80 1m " ‘00 " M tim " 10” “III
INPUT (IIIV INS) FIEnUENCV (hr)
Spot Noise Voltage " Spot Noise Current vs
Frequency Frequency
"0'3! VOLTAGE (WI
NOISE CURRENT (M/V'Ffi)
10 1” " ttm III 100 " ttlt
FMUEICY (Hz) FREQUENCY (HI)
PHASE (MGIEES)
Turn On Delay vs
Component Values and Gain
ll .1 .23 1.5.6 .1 I 31.01.11,?
IUBN-ON TIME DELAY (SECONDS)
Channel Separation vs
Frequency
cmcun or FIGUIE t
ttt 5010020050qu tt 5k10k20l
mzouzucv (W)
Output Amplifier Open Loop
Gain and Phase vs
Frequency
GAIN (dl)
PHASE (DEGREES)
Ill IN " 10k INk " WI 1MB
rnsuuzucv (Hz)
Input Amplifier
DC Output Voltage vs
Temperature (Pins 3, 14)
7‘ f" f"
9 on a
HUMP DC OUYFUT VOLTAUE IV)
-SO -N t) " 50 15 too
TEMPEIAIUIE ('C)
TL/H/7094-10
gif Natiqnal _
Semiconductor
LM383/LM383A 7 Watt Audio Power Amplifier
General Description
The LM383 is a cost effective, high power amplifier suited
for automotive applications. High current capability (3.5A)
enables the device to drive low impedance loads with low
distortion. The LM383 is current limited and thermally pro-
tected. High voltage protection is available (LM383A) which
enables the amplifier to withstand 40V transients on its sup-
ply. The LM383 comes in a 5-pin TO-220 package.
Features
I! High peak current capability (3.5A)
Large output voltage swing
Externally programmable gain
Wide supply voltage range (5V-20V)
Few external parts required
Low distortion
High input impedance
No turn-on transients
High voltage protection available (LM383A)
Low noise
AC short circuit protected
Equivalent Schematic
F EOVS
o Vnur
HNPUT -tre0T
Connection Diagram
Plastic Package
5 SUPPLY VOLTAGE
4 OUTPUT
3 GROUND
2 INVERTING INPUT
I NON-IHVERTING INPUT
Order Number LM383T or LM383AT
See NS Package Number T058
TL/H/7145-2
TL/H/7145-1
V888W1/888W'I
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This file is the datasheet for the following electronic components:
LM1897N - product/lm1897n?HQS=TI-nu|I-nu|I-dscatalog-df-pf-null-wwe
