ADP3810AR-4.2-ADP3810AR-8.4 Fast Delivery |IC PHOENIX CO.,LIMITED

ADP3810AR-4.2 ,0.4-18V; 500mW; secondary, off-line battery changer controller. For battery changer controlled for: lilon batteries (ADP3810) NiCad, NiMH batteries (ADP3811)SPECIFICATIONS (–408C ≤ T ≤ +858C, V = 10.0 V, unless otherwise noted)A CCADP3810Parameter Conditio ..
ADP3810AR-42 ,Secondary Side, Off-Line Battery Charger ControllersSpecifications subject to change without notice.REV. 0–2–ADP3810/ADP3811ABSOLUTE MAXIMUM RATINGS PI ..
ADP3810AR-8.4 ,0.4-18V; 500mW; secondary, off-line battery changer controller. For battery changer controlled for: lilon batteries (ADP3810) NiCad, NiMH batteries (ADP3811)Specifications subject to change without notice.REV. 0–2–ADP3810/ADP3811ABSOLUTE MAXIMUM RATINGS PI ..
ADP3811AR ,Secondary Side, Off-Line Battery Charger ControllersSPECIFICATIONS (–408C ≤ T ≤ +858C, V = 10.0 V, unless otherwise noted)A CCADP3810Parameter Conditio ..
ADP3820AR-4.1 ,±1% Precision, Single Cell Li-Ion Battery ChargerSpecifications subject to change without notice.ORDERING GUIDEABSOLUTE MAXIMUM RATINGS*Input Voltag ..
ADP3820AR-4.2 ,±1% Precision, Single Cell Li-Ion Battery ChargerLithium-IonaBattery ChargerADP3820FUNCTIONAL BLOCK DIAGRAM
AK4706VQ , 2ch 24bit DAC with AV Switch & HD/SD Video Filter
AK4708EQ , AV SCART Switch with Two RGB Outputs
AK520S , ISDN SINGLE CHIP NT1/(U&S/T-INTERFACE) TRANSCEIVER
AK5354VT , Low Power 20bit ΔΣ ADC with PGA
AK5354VT , Low Power 20bit ΔΣ ADC with PGA
AK5359ET , 24-Bit 192kHz ΔΣ ADC

ADP3810AR-4.2-ADP3810AR-8.4
0.4-18V; 500mW; secondary, off-line battery changer controller. For battery changer controlled for: lilon batteries (ADP3810) NiCad, NiMH batteries (ADP3811)
REV.0
Secondary Side, Off-Line
Battery Charger Controllers
FUNCTIONAL BLOCK DIAGRAM
VSENSEVCS
COMP
OUT
VCCVREFGND
VCTRL
FEATURES
Programmable Charge Current
High Precision Battery Voltage Limit
Precision 2.000 V Reference
Low Voltage Drop Current Sense: 300 mV Full Scale
Full Operation in Shorted and Open Battery Conditions
Drives Diode-Side of Optocoupler
Wide Operating Supply Range: 2.7 V to 16 V
Undervoltage Lockout
SO-8 Package
ADP3810
Internal Precision Voltage Divider for Battery Sense
Four Final Battery Voltage Options Available: 4.2 V,
8.4 V, 12.6 V, 16.8 V
ADP3811
Adjustable Final Battery Voltage
APPLICATIONS
Battery Charger Controller for:
LiIon Batteries (ADP3810)
NiCad, NiMH Batteries (ADP3811)
GENERAL DESCRIPTION
The ADP3810 and ADP3811 combine a programmable current
limit with a battery voltage limit to provide a constant current,
constant voltage battery charger controller. In secondary side,
off-line applications, the output directly drives the diode side of
an optocoupler to give isolated feedback control of a primary
side PWM. The circuitry includes two gain (gm) stages, a preci-
sion 2.0 V reference, a control input buffer, an Undervoltage
Lock Out (UVLO) comparator, an output buffer and an over-
voltage comparator.
The current limit amplifier senses the voltage drop across an
external sense resistor to control the average current for charg-
ing a battery. The voltage drop can be adjusted from 25 mV
to 300 mV, giving a charging current limit from 100 mA to
1.2 amps with a 0.25 Ω sense resistor. An external dc voltage
on the VCTRL input sets the voltage drop. Because this input
is high impedance, a filtered PWM output can be used to set
the voltage.
As the battery voltage approaches its voltage limit, the voltage
sense amplifier takes over to maintain a constant battery volt-
age. The two amplifiers essentially operate in an “OR” fash-
ion. Either the current is limited, or the voltage is limited.
The ADP3810 has internal thin-film resistors that are trimmed
to provide a precise final voltage for LiIon batteries. Four volt-
age options are available, corresponding to 1-4 LiIon cells as
follows: 4.2 V, 8.4 V, 12.6 V and 16.8 V.
The ADP3811 omits these resistors allowing any battery volt-
age to be programmed with external resistors.
ADP3810/ADP3811–SPECIFICATIONS
(–408C ≤ TA ≤ +858C, VCC = 10.0 V, unless otherwise noted)
VOLTAGE SENSE
REFERENCE
OUTPUT
UNDERVOLTAGE LOCKOUT
POWER SUPPLY
OVERVOLTAGE COMPARATOR
NOTES20 kΩ resistor from current sense voltage to VCS pin.Applies to 4.2 V, 8.4 V, 12.6 V and 16.8 V options. Includes all error from offset voltage, bias current, resistor divider and voltage reference.Does not include attenuation of input resistor divider for ADP3810.0.1 μF load capacitor required for reference operation.Full scale is the programmed final battery voltage: 4.2 V, 8.4 V, 12.6 V or 16.8 V for the ADP3810 or 2.0 V at VSENSE for the ADP3811.
All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods.
Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . .–0.4 V to 18 V
VCTRL, VCS Input Range . . . . . . . . . . . . . . . . . .–0.4 V to VCC
VSENSE Input Range (ADP3811) . . . . . . . . . . . .–0.4 V to VCC
VSENSE Input Range (ADP3810) . . . . . . . . . . .–0.4 V to 20 V
Maximum Power Dissipation . . . . . . . . . . . . . . . . . .500 mW
Operating Temperature Range . . . . . . . . . . .–40°C to +85°C
Storage Temperature Range . . . . . . . . . . . . .–65°C to 150°C
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . .+300°C
ORDERING GUIDE
WARNING!
ESD SENSITIVE DEVICE
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP3810/ADP3811 features proprietary ESD protection circuitry, permanent
damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper
ESD precautions are recommended to avoid performance degradation or loss of functionality.
PIN DESCRIPTION
PIN CONFIGURATION
VSENSE
VCS
COMP
OUT
VCC
VREF
GND
VCTRL
BATTERY
VIN
Figure 1.Simplified Battery Charger
ADP3810/ADP3811
TEMPERATURE – °C
REFERENCE VOLTAGE – Volts
2.004
Figure 2.Reference Output Voltage
vs. Temperature for Two Typical Parts
FREQUENCY – Hz
PSRR – dB
1001k1M10k100k
–40
Figure 5.Reference PSRR vs.
Frequency
TEMPERATURE – °C
CURRENT SENSE VOLTAGE – mV
–302
Figure 8.Full-Scale Current Sense
Voltage vs. Temperature
LOAD CURRENT – mA
DROPOUT VOLTAGE – mV
100
Figure 3.Reference Drop-Out Volt
age (VCC–VREF) vs. Load Current
FREQUENCY – Hz
REFERENCE NOISE DENSITY – nV/
25001010k1001k
2000
Figure 6.Reference Noise Density
vs. Frequency
SUPPLY VOLTAGE, VCC – Volts
CURRENT SENSE VOLTAGE – mV
–302
Figure 9.Full-Scale Current Sense
Voltage vs. VCC
Figure 4.Reference Dropout Voltage
vs. Temperature
Figure 7.Charge Current vs. Control
Voltage
FREQUENCY – Hz
OPEN-LOOP GAIN – dB
–401M10k
1001k
100k
PHASE SHIFT – Degrees
Figure 10.GM1 Open-Loop Gain and
Phase vs. Frequency
–Typical Performance Characteristics
FREQUENCY – Hz
OPEN-LOOP GAIN – dB
–401M10k
1001k
100k
PHASE SHIFT – Degrees
Figure 11.GM2 Open-Loop Gain and
Phase vs. Frequency
TEMPERATURE – °C
GM2 OFFSET – mV
–1.0
Figure 14.ADP3811 GM2 Offset vs.
Temperature
SUPPLY VOLTAGE, VCC – Volts
SENSE
BIAS CURRENT – nA
0.5
Figure 17.ADP3811 VSENSE Bias
Current vs. VCC
TEMPERATURE – °C
VOLTAGE SENSE ACCURACY – %
–1.0
Figure 12.ADP3810 Voltage Sense
Accuracy vs. Temperature
SUPPLY VOLTAGE, VCC – Volts
GM2 OFFSET – mV
–1.0
Figure 15.ADP3811, GM2 Offset
vs. VCC
VOV% – %
QUANTITY – Parts
100
Figure 18.Overvoltage Comparator
Distribution (VOV%)
Figure 13.ADP3810 Voltage Sense
Accuracy vs. VCC
Figure 16.ADP3811 VSENSE Bias
Current vs. Temperature
Figure 19.Overvoltage Comparator
Threshold (VOV%) vs. Temperature
ADP3810/ADP3811
OUTPUT GAIN (VOUT/VCOMP) – V/V
QUANTITY – Parts
160
Figure 20.Output Gain (VOUT/VCOMP)
Distribution
APPLICATIONS SECTION
Functional Description
The ADP3810 and ADP3811 are designed for charging NiCad,
NiMH and LiIon batteries. Both parts provide accurate voltage
sense and current sense circuitry to control the charge current
and final battery voltage. Figure 1 shows a simplified battery
charging circuit with the ADP3810/ADP3811 controlling an
external dc-dc converter. The converter can be one of many
different types such as a Buck converter, Flyback converter or a
linear regulator. In all cases, the ADP3810/ADP3811 maintains
accurate control of the current and voltage loops, enabling the
use of a low cost, industry standard dc-dc converter without
compromising system performance. Detailed realizations of
complete circuits including the dc-dc converter are included
later in this data sheet.
The ADP3810 and ADP3811 contain the following blocks
(shown in Figure 1):Two “GM” type error amplifiers control the current loop
(GM1) and the voltage loop (GM2).A common COMP node is shared by both GM amplifiers
such that an RC network at this node helps compensate both
control loops.A precision 2.0 V reference is used internally and is available
externally for use by other circuitry. The 0.1 μF bypass ca-
pacitor shown is required for stability.A current limited buffer stage (GM3) provides a current out-
put, IOUT, to control an external dc-dc converter. This out-
put can directly drive an optocoupler in isolated converter
applications. The dc-dc converter must have a control scheme
such that higher IOUT results in lower duty cycle. If this is
not the case, a simple, single transistor inverter can be used
for control phase inversion.An amplifier buffers the charge current programming volt-
age, VCTRL, to provide a high impedance input.An UVLO circuit shuts down the GM amplifiers and the
output when the supply voltage (VCC) falls below 2.7 V. This
protects the charging system from indeterminate operation.A transient overshoot comparator quickly increases IOUT
Description of Battery Charging Operation
The IC based system shown in Figure 1 charges a battery with a
dc current supplied by a dc-dc converter, which is most likely a
switching type supply but could also be a linear supply where
feasible. The value of the charge current is controlled by the
feedback loop comprised of RCS, R3, GM1, the external dc-dc
converter and a dc voltage at the VCTRL input. The actual
charge current is set by the voltage, VCTRL, and is dependent
upon the choice for the values of RCS and R3 according to the
formula below:
ICHARGE=1
RCS×R3
80kΩ×VCTRL
Typical values are RCS = 0.25 Ω and R3 = 20 kΩ, which result
in a charge current of 1.0 A for a control voltage of 1.0 V. The
80 kΩ resistor is internal to the IC, and it is trimmed to its ab-
solute value. The positive input of GM1 is referenced to
ground, forcing the VCS pin to a virtual ground.
The resistor RCS converts the charge current into the voltage at
VRCS, and it is this voltage that GM1 is regulating. The voltage
at VRCS is equal to –(R3/80 kΩ) VCTRL. When VCTRL equals
1.0 V, VRCS equals –250 mV. If VRCS falls below its pro-
grammed level (i.e., the charge current increases), the negative
input of GM1 goes slightly below ground. This causes the out-
put of GM1 to source more current and drive the COMP node
high, which forces the current, IOUT, to increase. A higher IOUT
decreases the drive to the dc-dc converter, reducing the charg-
ing current and balancing the feedback loop.
As the battery approaches its final charge voltage, the voltage
loop takes over. The system becomes a voltage source, floating
the battery at constant voltage thereby preventing overcharging.
The constant voltage feature also protects the circuitry that is
actually powered by the battery from overvoltage if the battery is
removed. The voltage loop is comprised of R1, R2, GM2 and
the dc-dc converter. The final battery voltage is simply set by
the ratio of R1 and R2 according to the following equation
(VREF = 2.000 V):
VBAT=2.000V×R1+1
VCC – Volts
OUT
COMP
– V/V0318691215
Figure 21.Output Gain (VOUT/VCOMP)
vs. VCC
Figure 22.VSAT vs. Temperature
current loop, the higher IOUT reduces the duty cycle of the dc-dc
converter and causes the battery voltage to fall, balancing the
feedback loop.
Each GM stage is designed to be asymmetrical so that each am-
plifier can only source current. The outputs are tied together at
the COMP node and loaded with an internal constant current
sink of approximately 100 μA. Whichever amplifier sources
more current controls the voltage at the COMP node and there-
fore controls the feedback. This scheme is a realization of an
analog “OR” function where GM1 or GM2 has control of the
dc-dc converter and the charging circuitry. Whenever the cir-
cuit is in full current limiting or full voltage limiting, the respec-
tive GM stage sources an identical amount of current to the
fixed current sink. The other GM stage sources zero current
and is out of the loop. In the transition region, both GM stages
source some of the current to comprise the full amount of the
current sink. The high gains of GM1 and GM2 ensure a
smooth but sharp transition from current control to voltage con-
trol. Figure 24 shows a graph of the transition from current to
voltage mode, that was measured on the circuit in Figure 23 as
detailed below. Notice that the current stays at its full pro-
grammed level until the battery is within 200 mV of the final pro-
grammed voltage (10 V in this case), which maintains fast
charging through almost all of the battery voltage range. This
improves the speed of charging compared to a scheme that re-
duces the current at lower battery voltages.
The second element in a battery charging system is some form
of a dc-dc converter. To achieve high efficiency, the dc-dc con-
verter can be an isolated off-line switching power supply, or it
can be an isolated or nonisolated Buck or other type of switch-
ing power supply. For lower efficiency requirements, a linear
regulator fed from a wall adapter can be used. In the above dis-
cussion, the current, IOUT, controls the duty cycle of a switching
supply; but in the case of the linear regulator, IOUT controls the
pass transistor drive. Examples of these topologies are shown
later in this data sheet. If an off-line supply such as a flyback
converter is used, and isolation between the control logic and
the ADP3810/ADP3811 is required, an optocoupler can be in-
serted between the ADP3810/ADP3811 output and the control
input of the primary side PWM.
Charge Termination
If the system is charging a LiIon battery, the main criteria to de-
termine charge termination is the absolute battery voltage. The
ADP3810, with its accurate reference and internal resistors, ac-
complishes this task. The ADP3810’s guaranteed accuracy
specification of ±1% of the final battery voltage ensures that a
LiIon battery will not be overcharged. This is especially impor-
tant with LiIon batteries because overcharging can lead to cata-
strophic failure. It is also important to insure that the battery be
charged to a voltage equal to its optimal final voltage (typically
4.2 V per cell). Stopping at less than 1% of full-scale results in
a battery that has not been charged to its full mAh capacity,
reducing the battery’s run time and the end equipment’s operat-
ing time.
The ADP3810/ADP3811 does not include circuitry to detect
charge termination criteria such as –ΔV/Δt or ΔT/Δt, which are
program the VCTRL input to set the charge current. The high
impedance of VCTRL enables the inclusion of an RC filter to in-
tegrate a PWM output into a dc control voltage.
Compensation
The voltage and current loops have significantly different natu-
ral and crossover frequencies in a battery charger application, so
the two loops most likely need different pole/zero feedback com-
pensation. Figure 1 shows a single RC network from the
COMP node to ground. This is primarily for low frequency
compensation (fC< 100 Hz) of the voltage loop. Since the
COMP node is shared by both GM stages, this compensation
also affects the current loop. The internal 200 Ω resistor does
change the zero location of the compensation for the current
loop with respect to the voltage loop. To provide a separate
higher frequency compensation (fC ~ 1 kHz–10 kHz), a second
series RC may be needed. A detailed calculation of the com-
pensation values is given later in this data sheet.
ADP3810 and ADP3811 Differences
The main difference between the ADP3810 and the ADP3811
is illustrated in Figure 1. The resistors R1 and R2 are external
for the ADP3811 and internal for the ADP3810. The ADP3810
is specifically designed for LiIon battery charging, and thus, the
internal resistors are precision thin-film resistors laser trimmed
for LiIon cell voltages. Four different final voltage options are
available in the ADP3810: 4.2 V, 8.4 V, 12.6 V, and 16.8 V.
For slightly different voltages to accommodate different LiIon
chemistries, please contact the factory. The ADP3811 does not in-
clude the internal resistors, allowing the designer to choose any
final battery voltage by appropriately selecting the external resis-
tors. Because the ADP3810 is specifically for LiIon batteries,
the reference is trimmed to a tighter accuracy specification of
±1% instead of ±2% for the ADP3811.
VCTRL Input and Charge Current Programming Range
The voltage on the VCTRL input determines the charge current
level. This input is buffered by an internal single supply ampli-
fier (labeled BUFFER) to allow easy programmability of VCTRL.
For example, for a fixed charge current, VCTRL can be set by a
resistor divider from the reference output. If a microcontroller
is setting the charge current, a simple RC filter on VCTRL enables
the voltage to be set by a PWM output from the micro. Of
course, a digital-to-analog converter could also be used, but the
high impedance input makes a PWM output the economical
choice. The bias current of VCTRL is typically 25 nA, which
flows out of the pin.
The guaranteed input voltage range of the buffer is from 0.0 V
to 1.2 V. When VCTRL is in the range of 0.0 V to 0.1 V, the out-
put of the internal amplifier is fixed at 0.1 V. This corresponds
to a charge current of 100 mA for RCS = 0.25 Ω, R3 = 20 kΩ.
The graph of charge current versus VCTRL in Figure 7 shows this
relationship. Figure 1 shows a diode in series with the buffer’s
output and a 1.5 MΩ resistor from VREF to this output. The
diode prevents the amplifier from sinking current, so for small
input voltages the buffer has an open output. The 1.5 MΩ
resistor forms a divider with the internal 80 kΩ resistor to fix the
output at 0.1 V, i.e., about 10% of the maximum current. This
corresponds to the typical trickle charge current level for NiCad
ADP3810/ADP3811
charge current levels can be obtained by either reducing the
value of RCS or increasing the value of R3. The main penalty of
increasing R3 is lower efficiency due to the larger voltage drop
across RCS, and the penalty of decreasing RCS is lower accuracy
(but higher efficiency) as discussed below.
VREF Output
The internal band gap reference is not only used internally for
the voltage and current loops, but it is also available externally if
an accurate voltage is needed. The reference employs a pnp
output transistor for low dropout operation. Figure 3 shows a
typical graph of dropout voltage versus load current. The refer-
ence is guaranteed to source 5 mA with a dropout voltage of
400 mV or less. The 0.1 μF capacitor on the reference pin is in-
tegral in the compensation of the reference and is therefore re-
quired for stable operation. If desired, a larger value of capacitance
can also be used for the application, but a smaller value should
not be used. This capacitor should be located close to the VREF
pin. Additional reference performance graphs are shown in Fig-
ures 2 through 6.
Output Stage
The output stage performs two important functions. It is a
buffer for the compensation node, and as such, it has a high im-
pedance input. It is also a GM stage. The OUT pin is a current
output to enable the direct drive of an optocoupler for isolated
applications. The gain from the COMP node to the OUT pin is
approximately 5 mA/V. With a load resistor of 1 kΩ, the voltage
gain is equal to five as specified in the data sheet. A different
load resistor results in a gain equal to RL × (5 mA/V). Figures
20 and 21 show how the gain varies from part to part and versus
the supply voltage, respectively. The guaranteed output current
is 5 mA, which is much more than the typical 1 mA to 2 mA re-
quired in most applications.
Current Loop Accuracy Considerations
The accuracy of the current loop is dependent on several factors
such as the offset of GM1, the offset of the VCTRL buffer, the ra-
tio of the internal 80 kΩ compared to the external 20 kΩ resis-
tor, and the accuracy of RCS. The specification for current loop
accuracy states that the full-scale current sense voltage, VRCS, of
–300 mV is guaranteed to be within 15 mV of this value. This
assumes an exact 20 kΩ resistor for R3. Any errors in this resis-
tor will result in further errors in the charge current value. For
example, a 5% error in resistor value will add a 5% error to the
charge current. The same is true for RCS, the current sense resis-
tor. Thus, 1% or better resistors are recommended.
As mentioned above, decreasing the value of RCS increases the
charge current. Since it is VRCS that is specified, the actual
value of RCS is not accounted for in the specification. An example
where RCS = 0.1 Ω illustrates its impact on the accuracy of the
charge current. The range of VRCS is from –25 mV ± 5 mV to
–300 mV ± 15 mV. This results in a charge current range from
250 mA ± 50 mA to 3 A ± 150 mA, as opposed to a charge cur-
rent range of 100 mA ± 20 mA to 1.2 A ± 60 mA for RCS =
0.25 Ω. Thus, not only is the minimum current changed, but
the absolute variation around the set point is increased (although
the percentage variation is the same).
Voltage Loop Accuracy Considerations
the ADP3810 is specified with respect to the final battery volt-
age. This is tested in a full feedback loop so that the single ac-
curacy specification given in the specification table accounts for
all of the errors mentioned above. For the ADP3811, the resis-
tors are external, so the final voltage accuracy needs to be deter-
mined by the designer. Certainly, the tolerance of the resistors
has a large impact on the final voltage accuracy, and 1% or bet-
ter is recommended.
Supply Range
The supply range is specified from 2.7 V to 16 V. However, a
final battery voltage option for the ADP3810 is 16.8 V. The
16.8 V is divided down by the thin film resistors to 2.0 V inter-
nally. Thus, the input to GM2 never sees much more than 2.0 V,
which is well below the VCC voltage limit. In fact, VCC can be
fixed to 2.7 V and the ADP3810 will still control the charging of
a 16.8 V battery stack. The ADP3811, with external resistors,
can charge batteries to voltages well in excess of its supply volt-
age. However, if the final battery voltage is above 16 V, VCC
cannot be supplied directly from the battery as it is in Figure 1.
Alternative circuits must be employed as will be discussed later.
Decoupling capacitors should be located close to the supply pin.
The actual value of the capacitors depends on the application,
but at the very least a 0.1 μF capacitor should be used.
OFF-LINE, ISOLATED, FLYBACK BATTERY CHARGER
The ADP3810 and ADP3811 are ideal for use in isolated charg-
ers. Because the output stage can directly drive an optocoupler,
feedback of the control signal across an isolation barrier is a
simple task. Figure 23 shows a complete flyback battery charger
with isolation provided by the flyback transformer and the
optocoupler. The essential operation of the circuit is not much
different from the simplified circuit described in Figure 1. The
GM1 loop controls the charge current, and the GM2 loop con-
trols the final battery voltage. The dc-dc converter block is
comprised of a primary side PWM circuit and flyback trans-
former, and the control signal passes through the optocoupler.
The circuit in Figure 23 incorporates all of the features neces-
sary to assure long battery life with rapid charging capability.
By using the ADP3810 for charging LiIon batteries, or the
ADP3811 for NiCad and NiMH batteries, component count is
minimized, reducing system cost and complexity. With the cir-
cuit as presented or with its many possible variations, designers
no longer need to compromise charging performance and bat-
tery life to achieve a cost effective system.
Primary Side Considerations
A typical current-mode flyback PWM controller was chosen for
the primary control circuit for several reasons. First and most
importantly, it is capable of operating from very small duty
cycles to near the maximum designed duty cycle. This makes it
a good choice for a wide input ac supply voltage variation re-
quirement, which is usually between 70 V–270 V ac for world
wide applications. Add to that the additional requirement of
0% to 100% current control, and the PWM duty cycle must
have a wide range. This charger achieves these ranges while
maintaining stable feedback loops.
The detailed operation and design of the primary side PWM is

Partno	Mfg	Dc	Qty	Available	Descript
ADP3810AR-4.2 \|ADP3810AR42	AD ?	N/a	80	avai	0.4-18V; 500mW; secondary, off-line battery changer controller. For battery changer controlled for: lilon batteries (ADP3810) NiCad, NiMH batteries (ADP3811)
ADP3810AR-8.4 \|ADP3810AR84	ADI	N/a	20	avai	0.4-18V; 500mW; secondary, off-line battery changer controller. For battery changer controlled for: lilon batteries (ADP3810) NiCad, NiMH batteries (ADP3811)