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ADS7886SDCKT ,2.35V-5.25V, 12 bit, 1MSPS, Serial ADC 6-SC70 -40 to 125Features 3 DescriptionThe ADS7886 is a 12-bit, 1-MSPS analog-to-digital1• 1-MHz Sample Rate Serial ..
AD9610BH
Wide Bandwidth, Fast Settling Operational Amplifier
ANALOG
DEVICES
Wide Bandwidth, Fast Settling
Operational Amplifier
A0961!)
FEATURES
Ultrastable Unity Gain Bandwidth (100MHz)
B'Endwidth Is Independent of Gain Settings
18ns Settling to 0.1%
Low Power Dissipation (630an)
Complete Overdrive Protection
Low Distortion ITHD: -SSdBe a 20MHz,
-78dBc @ SMHz, -100dBc © 10kHz)
Excellent DC Specifieations
Available Processed to MlL-STD 883
APPLICATIONS
Driving Flash Converters
High Speed DAC IN Converters
Radar, IF Processors
Broadband, Digital Radio
Photodlode Preamps (FUR)
ATE/Pulse Generators
lmaging/Display Drivers
GENERAL DESCRIPTION
The AD9610 is a fast settling, wide bandwidth, dc coupled,
operational amplifier which combines superior dc specifications
and exceptional dynamic performance with impeccable spectral
purity (harmonic distortion, intermodulation distortion, noise,
etc.) over the full bandwidth. This combination provides re-
markable versatility and utility for high speed designers.
Thin-film technology and innovative design techniques help
assure stable operation over the complete operating temperature
range. Input offset voltage is t0.3rnV, with 5wVf'C drift; input
bias currents are t15WA with t30nh/"C drift.
Unique internal architecture employing current feedback keeps
the AD9610 inherently stable over its complete gain range and
assures wide bandwidth at all gain settings. With G-- - l, -3dB
bandwidth is 120MHz; with G = - 10, - 3dB bandwidth is
100MHz. When G= -50, the --3dB bandwidth is 60MHz.
Slew rate, fall time and settling time are also independent of
F tequency domain performance for the AD9610 is unmatched.
The part can be used in applications requiring wide spurious
free dynamic range. At 10kH2 total harmonic distortion (THD)
is -100dBc; at IMH2 the THD is -8$dBc; at 20MHz the
THD is. - 59dBc. Third order intermodulation distortion is
similarly impressive, which is often required in communications
applications.
FUNCTIONAL BLOCK DIAGRAM
"s, 's, - Fs,
dB - \\ \GT
, N, "N
so 60 " " " 100
0 " 20 30 IO
AD9610 inverting Gain
The design of the AD9610 makes it easy to apply. The unit
requires no external compensation. An internal 1.Skft feedback
resistor is available to the user by connecting Pin 4 to Pin 11.
This resistor is trimmed for gain accuracy and should be used
when the full bandwidth of the amplifier is required. To achieve
higher gains, and for lower bandwidth applications, an external
resistor can be used. Pins 2 and 8 are bypass pins and should be
connected to ground through 33 - 500 resistors and 0.1WF
ceramic capacitors; effective decoupling of the power supplies is
also important to obtain optimum high frequency performance.
Two temperature ranges are available. The AD9610BH is
guaranteed over a case temperature range of - 25°C to + 85°C;
the AD9610TH is for a range of - 55°C to + 125°C. The AD9610
is available in versions compliant with MIL-STD-883. Refer to
the Analog Devices Military Products Databook or current AD9610/
883B data sheet for detailed specifications.
M881tl--SPEtyFliym0lG
in ELECTRICAL CHARACTERISTICS(5V= sltirA= --1tF,tu=1iitl0n;Rrr=16enititrGa)
ADNIOBH‘ AD96tttTH2
W193” Min/Max @ Min/Max tii) '
Typical
Parameter (Conditions) Iii) + 25NI -- 25°C + 25''C + 85''C - 55''C + 25°C + 125''C Units
J Ofrsetvoltage t0.3 t4.0 tl.0 :2.5 :4.0 tl.0 :2.5 mV
J Offset Voltage Tcs t5 , 25 t 25 wW'C
J Input Bias Current
{matting t.5 :56 :15 :35 :56 :15 35 wh
Noninverting t 15 t 75 t 50 t 62 t 75 t 50 + 62 wA
J Input Bias Current Tc' T
Inverting t 70 I 330 : 330 nA/°C
Noninverting t 30 t 200 t 200 nA/°C
f Noninverting
Impedance 200k n
Capacitance 2 pF
' thtnrnon-Modernput t5 t5 :5 t5 t5 :5 t5 V
J Intcmll Feedback Ruistbr (RF) 1500 1490/ 1490/
15 10 l S 10 n
' Rr Temperature Coefficient - 25 t 25 - 25 t 25 ppm/°C
J Common-Mode Rejection Ratio (CMRRf >50 >35 235 2 35 235 2 35 2 35 dB
CMRRtRr=t500n;Rm--x150mhvs--W) >60 dB
J Common-Modc Sensitivity (CMS),s .
Referred to Inpu1(AVs = 5V)
- CMS 3 8 8 8 8 8 8 wh/V
+ CMS 3 8 8 8 8 8 8 WA/ll
CMSVOLTAGE 62 250 250 250 250 250 250 dB
' Output Impedance (dc to 100kH2) 0.05 n
J otuputvoitageSwing(RuaAD--200n) :10 2:9 2:9 2:9 2:9 2:9 2:9 V
' OutputCumm t 50 2 :50 2 t 50 2 :50 2 t 50 2 :50 2 t 50 mA
(Continuous)
J Open LoopTransimpcdanceGain (2000 Load) >15 20.7 20.9 20.7 >0.7 20.9 >0.7 Mn
J Supply Currents 21 s27 sc25 <27 s27 s25 s27 m
PowerCtmsumption6 630 s810 $750 $810 5810 $750 5810 mW
J Power Supply Rejection Ratio (PSRR)' > 50 2 35 235 235 235 235 2 35 dB
PSRR(RF= 15000.;Rm: 1500;AV5= 10V) >60 dB
J Power Supply Sensitivity (PSS),7
Referred to Input (AVs = 10V)
PSSVOLTAGE 65 50 50 so so 50 50 dB
- PSS 3 8 8 8 8 8 8 [LAN
+ PSS 3 8 8 8 8 8 8 WA/V
ht ELECTRICAL thll1RhtTERlSTlt$:ev=. :15V;A.= -1tl;ltrt=150n;flr--1dil0lGa--ral0n)
Bandwidth ( - MB) (Vom- = 10thnV p-p)
J G= - 10 >100 280 280 280 280 280 ago MHz
Amplitude of Peaking:
J DCto60MHz 0 0.4 s0.2 $1.0 $0.4 s0.2 1.0 dB
' >60MHz 0 s0.6 50.3 sl.8 50.6 50.3 sl.8 dB
' Phase Nonlinearity (dc to 45MHi) l a
' Risc(Fall) Timc(Vou-r=5V Step) <35 s4 s4 s4.3 s4 -4 54.3 ns
' S1ewRate0lovr--i8Vstep) >35 23 23 >2.4 23 >3 22.4 kVIps
' Sett1ingTime to0.1% (G= -10;
5V Output Step) 18 529 s25 s29 s29 s25 s29 ns
f SettiingTime to 0.02% (G = - 10;
5V Output Step) 30 ns
' overshootAmplitudeNovra-- SVOutput Step) _ <4 s14 s8 $18 s14 s8 s18 %
I Propagation Delay 3.3 s4.0 $4.0 ss4.0 4.0 $4.0 54.0 ns
J Total Harmonic Distortion (Freq. = 20 MHz;
Output Voltage = 2V p-p) SS 50 50 50 50 50 50 dB
* Input Noise (RLOAD = 1000)
voltage(5MHito 150MHz) 0.7 51.2 51.5 52.0 51.2 51.5 52.0 i' nvn/E
Current (SMHz to 150MHz) 23 s29 s30 s35 $29 s30 s35 i pA/VE
LL-,-,,,-,,,
A0961!)
AD9610BHfrH AD96IOBH AD96lOTH
Sub- Typical Min/Max © Min/Max @
Group @+25'C -25°C +25''C +85''C -5!PC l +25''C +125°C Units
OTHER INFORMATION
Case to Ambient, 8mg 65 * * * * * * "C/W
(Still Air; No Heat Sink)
Case to Ambient, 6mg 38 * * * * * * "C/W
(500 LFPM Air; No Heat Sink)
MTBF' al.48 x lif * * * * r * hours
/ 100% tested (See Notes 1 and 2). .
' Specifications guannneed by design; not tested.
'Specification same as AD9610BH/TH typical specification.
'AD96103H parameters ptwcded by a check (J) are tested at + 25°C ambient temperature; performance is guaranteed over the
industrial temperature range t-25"C to + 85°C) case temperature.
'AD9610TH parameters preceded by a check (J) are tested at - 55°C case, + 25°C ambient, and + 125°C case temperatures.
Mil-processed versions are available.
3Otrset voltage Tc and bias current Tc are guaranteed over the respective temperature ranges.
'CMRR and PSRR apply only for mud conditions.
'CMS values can be used to determine the CMRR for specific gain settings according to the following worst case relationships:
R1 + SUPPLY
R2 r-et Vow
_ VSUPPLV
AVour = t-CMSI [Rd [Avsumy] ' [+CMS] IR21 F +%] [Avsumy] + [CMSvonl F /,'u,] [AVSUMY]
WHERE AVSUPPLY = A-Vsurrw AND A +Vsumy
CMRR = -20 LOG -lt.'-rs1.--
(1 + ii-P AVsumy)
'Supply current and power dissipation numbers are for quiescent operation (input is grounded). Values increase with
higher frequency operation.
Yss values can be used to determine the PSRR for stwcific gain settings according to the following worst case relationships
(See diagram in 5 above):
AV” = i-PSS (ttrl [AVsunul + (+Pss1 [R21 F + 'lu,) [AVsumv1 ' [Pssvou] F + J-,' ] [AVsumv]
WHERE AVMV = A-VWV OR A+Vsurrw
PSRR " -20 ura "r-st-rt'.'.-
'Recommended maximum inaction temperature is + 165'C, See Thermal Model.
'MTBF calculated using MIL-HNBK 217D; Ground Fixed; Temperature (case) = + 70°C.
Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS
Supply Voltages (t Vs) ................. 1 18V Power Dissipation ............. See Thermal Model
Operating Temperature Range (case) Junction Temperature ................. + 165°C
AD9610BH ................ - 25°C to + 85°C Storage Temperature Range ......... _ 65°C to + 150°C
AD9610TH/I‘H/883B ........... - 55°C to + 125''C Lead Temperature (soldering, 10 sec) ......... + 300°C
ORDERING GUIDE
Temperature Package Package
Model Range Description Option'
AD96IOBH - 25°C to + 85°C TO-S Style Metal Can H-12A
AD9610TH - 55°C to + 125°C TO-S Style Metal Can H-12A
AD9610TH/883B - 55°C to + 125°C TO-8 Style Metal Can H-12A
*For outline information see Package Information section.
Angelo '
LIFE TEST/BURN-IN CIRCUIT
M=RNCSSH2M1F
p=RN5507501F
1our=msooa1atzm
oaur=msou1o14ssa
THIS MICROCIRCUIT Is COVERED BY TECHNOLOGY GROUP (ll '3 MlL-M-38510
THEORY OF OPERATION
The advantages of the transimpedance AD9610 Operational
Amplifier become easier to understand when its operation is
compared to the operation of conventional high-speed op amps.
The operation of the AD9610 Operational Amplifier is similar
to a standard voltage-input differential amplifier in terms of
setting gain and calculating noise. The primary difference between
the two types is a Iow-impedance inverting input on the AD9610;
this causes the unit to use current feedback, rather than voltage
feedback, to achieve signal amplification.
Figure 1 and the discussion which follows help make a comparison
between the AD9610 and "conventional" devices.
Two equations are necessary to describe the amplifier shown in
Figure l.
Ihr: PAv-v‘v Vs
(,''s --0 van
Figure 1.
One equation is a rudimentary amplifier transfer function:
‘VOUT = A(as) vs (Equation A )
and the other sums the currents at the inverting ihput:
- =0 E tion B
SR_,N 'RC,' VSRF f qua )
Rearranging and reducing Equation B; and substituting from
Equation A results in a third equation:
Vow - - A(mmsRF/(RSRF + RINRF + RINRS)
(Equation C l
Vm 1 + A(w)RINRs/(RSRF + RNRP + RmRs)
For purposes of discussion, assume the amplifier shown in
Figure l exhibits a single-pole frequency response. When it
does, A(w)= Ao/(l +iurr) where Ao = open loop gain; and
l/r = the roll-off frequency. When these teims are substituted
into Equation c, the result is:
Vow = - AORSRF/(RSRF + RINRF + RWRS)
VI” 1 + ion + [AoRmRs/(RsR, +RINRF + RmRS>i
Based on the idea that
l + [AORmRs/(RSRF + RINRF + RINRS)]
is approximately equal to
AoRmRs/(RSRF + RINRF + RINRS)
and G (closed loop gain) = RF/Rm, it becomes possible to
simplify and substitute terms in the above equation to obtain:
Votrr = -G
1 + - + -
Ao Rr,,' k-s RF
The fundamental difference between the AD9610 and traditional
amplifiers becomes apparent at this point.
In traditional voltage-imput amplifiers, the input resistance (Rs)
approaches infmity. Consequently, l/Rs approaches zero; and
the term RF (l/RIN + I/Rs + l/RF) simplifies to the term
RF (l/Ro, + I/Rs). The latter can be reduced further to (G+ 1).
When substitutions are made, the gain/frequency relationship
for a traditional amplifier design is expressed as:
VOUT = -G
VIN 1+ it-'Tws+1,
There is a dramatically different result for the AD9610.
This difference is because the value of Rs in the transimpedance
amplifier is only 200. This is important when one realizes
Rs II Rm II RF; and Rs <<
(l/Rs + l/Rm + I/Rr) r-L-. l/Rs. Substituting terms, a direct
comparison with traditional amplifer relationships can be made:
Vow = --G
VIN ie E
1 + W, ]
Both amplifier types yield similar algebraic results, but there IS
one critical difference in how they are obtained.
As shown above, the closed loop gain (G) of the traditional
amplifier is multiplied by the frequency-dependent term of the
denominator; this means increasing frequencies or closed loop
gain accelerates the gain roll-off.
In the AD9610, however, the constant RF/Rs is multiplied by
the frequency-dependent term; this means bandwidth remains
relatively constant for any given value of gain.
A0961 0
Inside the AD9610, the design includes a 1.5k0 feedback resistor
to help reduce the effect of stray capacitances and make it easier
to apply the amplifier. This internal RF means the gain of the
AD9610 is set by varying Rm.
The differences in the architecture of the AD9610 vis-a-vis a
traditional op amp cause its closed-loop frequency response to
be considerably different from conventional units.
Figure 2 pictures a typical plot for a traditional single-pole
amplifier.
As shown, increasing the closed 1oo'p gain of a traditional op
amp decreases the bandwidth of the amplifier; the precise amount
GAIN = "
- 3dB FREQ. = MN:
GAIN = s
- 3dB FREQ " 20MHI
GAIN (d3)
FREQUENCV
Figure 2.
of change will be determined by the actual roll-off characteristics
of the op amp.
By contrast, the frequency response of. the AD9610 changes
very little when the gain is changed. Refer to Figure 3.
Variations in gain (established by varying values of Rm) have
only a negligible effect on the bandwidth of the amplifier.
(NOTE: For a more complete explanation of the mathematics involved
in comparing conventional op amps and the AD9610, refer to the
Analog Devices application note entitled "Using the AD9610 Trans-
impedance Amplifier.")
GAIN (d8)
FREQUENCY
Figure 3.
AD9610 FUNCTIONAL DESCRIPTION
Refer to Figure 4, AD9610 Functional Circuit.
GROUND
BYPASS
Figure 4. A09610 Functional Circuit
The most prominent characteristic illustrated in this model of
the unit is the combination of a high-impedance noninverting
terminal and a low-impedance inverting terminal. This is achieved
by buffering the noninverting terminal to create a high-impedance
input; while maintaining a low impedance through the 200
characteristic of the inverting input.
Because of the tow input impedance of the inverting input, all
of the input signal voltage is impressed across the input resistor
(Rm in Figure 6); this causes a direct voltage-to-current conversion
to take place.
Conventional op amps use a volts/volts transfer function,
while the transfer function of the AD9610 is volts/PA (or
resistance).
Signal current flowing in the inverting terminal (Pin 5) will flow
through the 200 resistor. The voltage developed across this
input impedance becomes the input signal for the internal
amplifier,
As a result of this action, the input current is converted to an
output voltage; this is the reason for the open loop transfer
function being expressed in ohms.
To compensate for variations in offset voltage and current in the
AD9610, both a voltage source and a current source are included
in the unit. Input offset voltage (Vos) is a dc error which appears
at the output as [Vos (1+ RF/RINH. In a similar fashion, the
input bias current (105) reflects as a dc error which appears at
the output as [log (ru)).
The current source connected to the inverting terminal effectively
models the input offset current; and although bias currents flow
in botlrterminais, the inverting input bias current is dominant.
The combined actions of the internal voltage and current sources
effectively compensate for discrepancies in offset voltage and
current.
Power supply voltages applied to the AD9610 are separated,
with one set of terminals designated for the output transistors
(Pins 10 and 12) and another set for the internal amplifier
(Pins 1 and 9). This splitting of the voltages makes it possible to
limit voltage swings and current at the output, and helps regulate
the junction temperatures of the output transistors.
