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AD8334ACPZ
Quad VGA with Ultralow Noise Preamplifier and Programmable RIN
F ig. 1 VGA-based
loop block diagram. V
VGA (LINEAR IN dB)
SIGNAL
rlll MDRM d.
)ESIGN AND (OPERATION
OF AUTOMATIC GAIN
CONTROL Loops
FOR RECEIVERS IN MODERN
(COMMUNICATIONS SYSTEMS
his article is intended to provide insight
ll into the effective operation of variable
gain amplifiers (VGA) in automatic gain
control (AGC) applications. Figure I is a gen-
eral block diagram for an AGC loop. The in-
put signal passes through the VGA to produce
the output level to be stabilized. The detec-
tor's output is compared against a setpoint
voltage to produce an error signal, which is
then integrated to produce a gain control volt-
age. This is applied to the control input of the
VGA. The attenuator shown be-
tween the VGA and the detec-
tor is used to align the maxi-
mum output level of the VGA
with the maximum input level
S lG NAL
O UTPUT
of the detector.
INPUT L I
RF ATTEN
cinteg Rstab
RSSI OUTPUT
i, DETECTOR
In the course of this article
several key issues will be ad-
dressed, including VGA types,
loop dynamics, detector types,
the operating level of VGA and
the operating level of the detec-
tor. Then an example applica-
INTEZ$¥OR I tion revolving around an
AD8367 VGA will be presented
SETPOINTINPUT for further discussion of the
material.
VGA TYPES
There are two major classes of VGA in use
today. The first is the so-called IVGA (input
VGA), which can be regarded as a passive
variable attenuator followed by a fixed-gain
amplifier. The second type is the output VGA
(OVGA), which is essentially equivalent to a
fixed-gain amplifier followed by a passive at-
tenuator.
An IVGA is the preferred choice for a re-
ceive AGC system because the available out-
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DANA WHITLOW
Analog Devices
Beaverton, OR
LOG LINBNV
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o 51015202530
TIME(ms)
A Fig. 2 Simulated response ofAGC loop
to large amplitude steps for various detectors.
put level at low distortion is relatively
independent of the gain setting. This
is the desired trait for an AGC sys-
tem, whose very object is to maintain
a constant output in the face of way-
ing input signal amplitude.
The OVGA is generally ill suited
to AGC applications because of its re-
duced output signal handling capabil-
ity at low gain settings and therefore
will not be discussed further here.
When a single IVGA is used in a
situation in which the VGA sets the
system noise floor, the output SNR is
essentially independent of the input
signal; it does not improve as is often
preferred. Occasionally, it is desirable
to cascade two VGAs in order to ame-
liorate this behavior or simply to ob-
tain more gain control range. Doing
so requires proper coordination of
the gain control inputs of the two de-
Vlces.
If the gain control of only the sec-
ond stage VGA is manipulated in the
weak signal regime, the signal level to
the first stage VGA's amplifier in-
creases with increasing input level, so
the output SNR improves with in-
creasing input level. It is necessary to
hand off the gain control from the
second stage to the first stage only
when overload of the first stage's alll-
plifier is imminent.
Alternatively, the two gain control
inputs may simply be driven in paral-
lel, in which case the output S/N (ex-
pressed in dB) improves at half the
rate at which the input level (also ex-
pressed in dB) rises. In cases where
the VGAs used have residual ripple in
their gain control functions, an addi-
tional benefit of this approach can be
obtained if the two gain control input
signals are intentionally offset by half
the period of the ripple. This can pro-
rlll MDRM d.
vide considerable reduction of the
ripple.
One of the benefits of using an
IVGA in an AGC loop is that the
VGA's gain control voltage bears an
accurate logarithmic relationship to
the input signal level when the loop is
in equilibrium. This means that the
gain control voltage may also be used
as an excellent received signal
strength indicator (RSSI).
LOOP DYNAMICS
Response time is an important is-
sue when designing any AGC loop.
There is usually a compromise be-
tween having the loop respond to un-
desired input level fluctuations as
rapidly as one would like, and having
it undesirably modify amplitude mod-
ulation on the signal. Additionally,
large and/or abrupt changes in the in-
put level may lead to unacceptable
recovery behavior, necessitating fur-
ther adjustments of the response
The issue of excessive loop band-
width deserves a bit more explana-
tion. If the loop responds too quickly,
it will introduce undesired gain mod-
ulation arising from the loop's efforts
to stabilize the output level of a signal
containing legitimate amplitude mod-
ulation. This is referred to as "gain
pumping." In the context of digital
modulation, the presence of apprecia-
ble gain pumping can result in signifi-
cant modulation errors and perhaps
even noticeable spectral re-growth in
extreme cases. A tolerable value of
gain pumping would generally be only
a fairly small fraction of 1 dB.
DETECTOR TYPES (DETECTOR
One convenient aspect of an AGC
loop is that the detector need not nec-
essarily have a very wide dynamic
range over which it obeys any particu-
lar law. This is because the detector
operates at a constant average level
when the AGC loop is in equilibrium;
thus, the detector should only need to
cope accurately with the level range
associated with a modulated signal.
However, as mentioned earlier, the
detector's response law (that is linear,
log, square law, etc.) can play a signifi-
cant role in determining the loop's dy-
namic response during large, abrupt
changes in signal level. Perhaps more
importantly, the detector’s response
law influences the dependency of the
loop's equilibrium level on the input's
waveform or crest factor.
F our detector types will be consid-
ered here: envelope detector; square-
law detector; true-RMS detector; and
log detector.
ENVELOPE DETECTOR
(RECTIFIER)
The output voltage of the enve-
lope detector is proportional to the
magnitude of the instantaneous RF
input voltage. Assuming that suffi-
cient low pass filtering is applied at
its output to eliminate RF ripple, this
detector produces a voltage propor-
tional to the envelope amplitude of
the RF signal.
Assuming that the loop's band-
width is made sufficiently small as to
avoid significant gain pumping, the
effect of the loop using an envelope
detector is to stabilize the average
rectified voltage of the signal. The re-
sulting power is therefore dependent
on the RF signal's envelope wave-
form. Such a loop acting on a con-
sunt-envelope signal such as GSM
will produce an average output power
which is different than that for a
heavily-amplitude-modulated signal,
such as CDMA or 64QAM.
The output of the envelope detec-
tor cannot go negative no matter how
weak the input signal, but may reach
extreme positive values in response to
very strong signals. Starting with the
AGC loop in equilibrium, a sudden
large increase in input amplitude
causes a very large initial increase in
detector output, which very rapidly
drives the loop towards lower gain.
On the other hand, an abrupt reduc-
tion of the input signal level (no mat-
ter by how many dB) cannot reduce
the detector output below zero, and
the loop's best response is to slew to-
wards equilibrium at a fairly low rate
until the detector output begins to
change by a significant fraction of the
reference voltage, at which point the
recovery trends towards an exponen-
tial decay. In the slew rate limited re-
gion, the gain of the signal path is
varying at a constant number of dB
per second.
Figure 2 shows the behavior of
such a loop for a large input level
step (note that curves for all four de-
tector types are superimposed on this
plot). These results were obtained
from simulations in which the VGA
has representative limits on the gain
range and on the maximum output
level. The detectors contrived for
these simulations have no particular
limits, on the grounds that in most
practical situations the designer will
scale the circuit so that the detector
does not limit appreciably before the
VGA does.
SQUARE-LAW DETECTOR
This type of detector has an in-
stantaneous output which is propor-
tional to the square of the instanta-
neous RF input voltage, which is
equivalent to saying that its output is
proportional to input power. This be-
havior, when incorporated into an
AGC loop of sensible bandwidth,
makes the loop's equilibrium average
output power independent of the in-
put waveform. As with the envelope
detector, the output can never go
negative, resulting in the loop having
a similar tendency towards slew rate
limited behavior when reacting to
abrupt decreases in input amplitude.
The response to large abrupt increas-
es in input amplitude can be even
more striking, however, because the
square-law detector characteristic ex-
aggerates the effect of the input in-
crease. The extent to which this hap-
pens depends on the clipping, level of
either the VGA or the detector,
whichever appears at a lower level.
TRUE-RMS DETECTOR
This detector comprises a square-
law detector followed by a low pass
filter followed by a square-root func-
tion. The low pass filter performs the
"mean" operation associated with the
root-mean-square (RMS) function,
and it should have a sufficiently long
time constant to smooth the output
variations of the squaring detector
that would otherwise arise from the
legitimate modulation of the signal.
Because of the square-root ele-
ment in this detector, the average
output is proportional to the signal
voltage, not power, so the loop's re-
sponse to small abrupt decreases or
increases of signal level should essen-
tially be the same as that for an enve-
lope detector, provided that the
added filter pole within the RMS de-
tector is correctly compensated for
elsewhere in the loop. The fact that
the added pole is located in a region
ll'lU MDRM d.
of the signal path that is square law
brings forth the possibility of the
large-step response being different
from that of the simple envelope de-
tector, which can indeed be seen in
the figure. Note that the RMS detec-
tor has a slightly slower recovery from
a large downward amplitude step
than does the standard envelope de-
tector, but a slightly faster recovery
(and a bit of overshoot) from a step
up in input amplitude. In common
with the square-law detector, the
true-RMS detector will make the
AGC loop's equilibrium point inde-
pendent of the RF signal waveform.
It should be noted that the pres-
ence of the long-tune-constant low
pass filter in this detector may have a
marked influence on loop dynamics;
indeed, this filter may even provide
the dominant pole in some designs.
This time constant must therefore be
coordinated with the remainder of
the loop design.
LOG DETECTOR
This type of detector produces an
output proportional to the logarithm
of the RF input voltage. Because this
behavior is complementary to that of
the linear-in-dB VGA in the loop, the
resulting loop dynamics are those of a
linear system, assuming that signal
level fluctuations during transients
remain within the measurement
range of the log detector. Subject to
that assumption, the AGC loop's re-
sponse to abrupt large changes in in-
put level will not be slew-rate limited,
and will often be faster to recover
from amplitude decreases.
As with the envelope detector, the
equilibrium point of an AGC loop us-
ing the log detector will depend on
the RF input waveform.
COMPARISON OF RESPONSES
WITH DIFFERENT DETECTORS
The AGC loops whose simulation
results are shown were designed so
that the small-signal response speeds
are identical. The results show that
the loop's large-step transient re-
sponse is markedly dependent on the
type of detector. At one extreme, the
log detector gives the fastest response
to large abrupt decreases in input lev-
el because the logarithmic curve has
a very steep slope for low inputs,
which exaggerates the loop's re-
sponse. However, the log detector
has a shallow slope for high input lev-
els, resulting in a diminished re-
sponse rate to sudden increases in
signal level. At the other extreme, the
square-law detector’s small slope near
zero input level gives it a very slug-
gish response to large decreases in in-
put amplitude. Conversely, the
square-law detector exaggerates the
response to large signals, giving the
fastest response to increasing signals.
The envelope and RMS detectors,
having intermediate characteristics,
give response speeds in between.
OPERATING LEVEL
OF DETECTOR
Ideally the operating level of the
detector should be set as high as pos-
sible in order to minimize the error
due to residual DC offsets. However,
other considerations often rule. For
signals with amplitude modulation,
the peak input to the detector when
the loop is in equilibrium must be no
higher than what the detector will
support, and so the average must be
lower. Even for constant-envelope
signals, the average level must be rea-
sonably lower than the maximum, so
that there is room for the detector
level to increase if the system input
level increases; otherwise, there
could be little or no error signal to
drive the loop back towards equilibri-
um. Note that there will generally be
unequal amounts of room for the de-
tector output to swing up or down
from the design equilibrium level,
which will make the apparent attack
and decay speeds of the loop differ.
DESIGN EXAMPLE
OF A WORKING AGC LOOP
Let's now put the above considera-
tions to work in a practical design.
The design assumptions and goals are
as follows:
. Signal modulation: W-CDMA (15
us/ers),. symbol rate = 3.84 Msym-
bols/s
. IFfrequency: 380 MHz
q VGA: AD8367
q Detector: AD8361 (true RMS, for
waveform independence)
. Power supply: 5 VDC
From these, reasonable constraints
will be established for operating lev-
els to maximize the adjacent channel
power ratio (ACPR) and AGC loop
bandwidth (to avoid excessive gain
pumping).
Previous bench measurements on
the AD8367 had revealed that
the best ACPR at 380 MHz occurs
with an output level of about 112
mVrms, which gives about -12, dBm
into 200 Q.
Detector Operating Level
The peak-to-average power ratio
for the chosen signal is about 18 dB.
When operating from a 5 V supply,
the maximum output level of the
AD8361 is about 4.8 V (from the
AD8361 datasheet). The squarer in
the detector will be assumed to go
into clipping at the same input level
that results in maximum output, for a
CW signal. Thus, assuming that the
peaks of the modulated signal should
not drive the detector's squarer into
clipping when the loop is in equilibri-
um, the average output level of the
detector must be such that it is at
least 18 dB below 4.8 V, 4.8 M 10-18/20
= 604 mV. Since the conversion gain
of the detector is 7.5 V/Vrms, the
loop-equilibrium level at the detec-
tor's input should be 604 mV/7.5 = 80
mVrms.
This level can be obtained from
the desired output level of the VGA
by adding a series resistor of 90 O,
which combines with the 225 9 input
resistance of the AD8361 to form a
voltage divider that achieves the de-
sired result. Note that this loads the
output of the VGA with 315 Q, which
means that the lowest additional par-
allel load impedance on the VGA
would be 547 Q in order to satisfy its
design minimum load impedance of
200 n. However, in this case more
than half of the VGA's power output
is going to be feeding the detector.
This could be remedied by driving
the VGA end of the 90 Q resistor
with an emitter follower, raising the
input impedance of the overall detec-
tor by the beta of the transistor used
in the follower. This would free up al-
most all the output current capability
of the VGA for use by the useful load.
Estimation of Target
AGC Loop Bandwidth
Here a judgment call must be
made, based on an empirical mea-
surement, to establish the maxi-
mum loop bandwidth that will avoid
intolerable gain pumping. For pur-
poses of this design example, it is
assumed that up to 0.5 dB p-p gain
rlll MDRM d.
variation is acceptable. An estimate
of the desired loop bandwidth will
be made by passing a W-CDMA sig-
nal through a spectrum analyzer
with very wide resolution band-
width, zero span and linear detec-
tor, and observe what video band-
width results in 0.5 dB p-p output
variation. The result turns out to be
200 Hz, which means the initial de-
sign of the AGC loop will have a
200 Hz bandwidth. The simulation
and measured results will be used
to see how this choice works out.
RMS Detector Filter
The RMS detector’s "mean value"
filter comprises an internal filter re-
sistance combined with an external
shunt capacitance. The effective val-
ue of the filter resistance varies with
drive level, from about 2000 Q at
very low drive level down to about
500 f2 at maximum drive level. For
this example a value of 1.8 kn will be
used, which was determined empiri-
cally for the operating level estab-
lished earlier.
In general, an AD8361 would be
taken into the lab with a W-CDMA
signal source in order to ascertain a
suitable filter capacitor value. Howev-
er, the previous measurement for loop
bandwidth gives a clue that allows a
reasonable estimate of the required
value to be made. A loop bandwidth
of 200 Hz resulted in a 0.5 dB p-p (--6
percent) variation of detector output.
It so happens that this is just about
the maximum amount of variation of
RMS filter level that still gives good
RMS accuracy. So, the bandwidth of
this filter will simply be made equal to
200 Hz, which requires a filter capaci-
tor of about 0.44 pF' against the 1.8
kn filter resistance.
Loop Dynamics Design
A first order loop will be devel-
oped, with a small-signal bandwidth
of 200 Hz. Note that the RMS detec-
tor's filter already contributes one
pole at 200 Hz, so the remainder of
the loop will clearly need to take this
into account. This will be achieved by
choosing Rmmp to create a zero at
200 Hz in conjunction with Cinteg
(see Appendix A). The response
speed of all the other elements in the
loop is so much faster than that of the
desired loop that all other poles can
be safely ignored.
The loop bandwidth designed will
apply only for small deviations from
the AGC loop's equilibrium level.
Large transients will behave differ-
ently because of the nonlinear char-
acter of the loop, and simulation
and/or breadboarding will be relied
upon to investigate the large signal
behavior.
The next items on the agenda are
the determinations of the incremen-
tal gains of the VGA and of the detec-
tor from the loop dynamics view-
point. For the VGA, this means the
slope of Vout versus control voltage,
not the RF gain.
VGA Gain
Vin and Vout will represent RMS val-
ues of the VGA's RF input and output,
respectively, and VA will represent the
gain control voltage". F rom examination
of the AD8367 performance data in the
datasheet for 240 MHz, combined with
a bit of extrapolation and rounding, 0
dB of gain is found to occur at a control
voltage of 0.1 V and the control slope is
50 dB/V. By formulating and then dif-
ferentiating Vout with respect to V, at
the equilibrium output of 112 1n\/:ns,
the incremental slope is evaluated to be
0.6447 V JV.
Next the Detector Slope
The nominal conversion gain of
the AD8361 is 7.5. However, a 90 f2
series resistor was added at the input
of the detector; this has the effect of
reducing the detector’s effective gain
to 5.357, which is the value that will
be used in the loop analysis.
Avoidance of Excessive
Recovery Delay
If the loop is left sitting with a very
low (or zero) input signal level for a
time, the output voltage of the inte-
grator will continue to rise until it
reaches saturation of the op-amp as
the loop tries to find more gain.
When a significant signal does sud-
denly arrive at the system input, one
has to wait for the integrator’s output
to ramp back down to 1 V before the
loop can begin reducing the gain. To
reduce this "overload delay" a 4.3:1
attenuator is inserted between the in-
tegrator and the control input of the
VGA so that the positive limit (nearly
5 V) of the integrator’s output is
about equal to the maximum effec-
tive control input (1 V) to the VGA.
rlll MDRM d.
o 4 30
0.3 20
:':'l 0.1 a
a o 3 o
u: -0.t .I /
... u:
-0.2 it -t0 f
-0.3 -20
o 2 q 6 8 101214161820 30
TlME(ms) -
A Fig. 3 The prototype circuit response to 400 2 4 6 8 IO " " 16 " 20
small amplitude steps is symmetrical and
shows exponential recovery.
Calculation of Component Values
In the loop, a net gain (excluding
the integrator for the moment) of 0.644
(VGA incremental slope) . 5.357 (ef-
fective detector slope)/4.3 (for the Vagc
atten) = 0.803, is obtained. For a loop
bandwidth of 200 Hz, the loop gain
should be unity at that frequency. If the
rest of the loop has a gain of 0.803, the
integrator must have a gain of 1.0/0.803
= 1.245, which requires that the reac-
tance of Chm," at 200 Hz be 1.245 times
the value of Rm. Mathematically, 1/(2 TT
. 200 . Cintog) = 1.245 . Bin; Rin . Cintog
= 639.2 us.
Now another constraint must be
also noted, which is that the AD8361
cannot source very much current,
TIME (ms)
A Fig. 4 The prototype circuit response to
large amplitude steps is asymmetrical and
exhibits slow-rate-limited recovery.
and can sink even less. Therefore, 50
k9 is chosen for Rin in order to mini-
mize total loading on the AD8361's
output, which leads to a value of
12.78 nF for Cinteg: Finally, in order
to compensate for the 200 Hz pole in
the RMS detector, Rcum is chosen as
62.3 kg to provide a loop zero at 200
Hz with Cinter. A 10 k9 pulldown is
also added on the AD8361's output to
improve its effective sinking capacity.
Lab Tests
A prototype circuit was construct-
ed according to the schematic in Ap-
pendix A. Figures 3 and 4 show the
-O.t o
RELATIVE GAIN (dB)
o 0.02 0.04 0.06 0.08 0.10
TlME(s)
A F ig. 5 The prototype circuit has about
0.2 dB p-p gain pumping.
prototype loop's response to small
and large (30 dB) input level steps,
respectively. Figure 5 shows the
measured gain pumping obtained by
capturing the signal at the gain-con-
trol input of the VGA with an oscillo-
scope and scaling into dB.
CONCLUSION
Numerous basic and finer points
of AGC design have been discussed,
and a detailed example of a practical
design was worked out. Practical con-
siderations and difficulties encoun-
tered along the way were empha-
sized, as opposed to reiterating text-
book loop design equations. Finally, a
working circuit was constructed and
measurement results presented. D
-u"-9 RF OUT (TO 50 n LOAD)
+5 v 160
AD8367
III I ICOM ICOM " III
2 " l I I 10 nF
ENBL HPFL l I .II
10 nF 3 1 thtrf ill-ir- , AD836t 8
RON I t INPT VPSI - VPOS SREF "-llt
4 " I " -- 2 7
MODE vpso - 10 nF T10 nF IREF VRMS -
91 0.44 oF
68 5 GAIN VOUT " I I W RFIN FLTR I
6 9 " Rt 4 5 C2
- DETO DCPL ts,-]. "f-' PWDN COMM :II
- 7 ICOM OCOM 8 M " U2
- Ut I "I',"
3.3 nF = = - 10 K
"f-ir- R4 33k 1an: 62K ill " -
W i I W
j, Ct R3
10 k Vrsst "I- R2
49.9 K
4 / ' w
- o , " 1 + 3 gt', OPERATING snpom'r
I . U3 --
APPENDIXA Il-ir- 22t0nF A0589 12v
SCHEMATIC OF BREADBOARD OF EXAMPLE 5 v "H-NW- II
CIRCUIT USING A TRUE-RMS DETECTOR. -- 10 K
= 3.3 K
