AD539JD ,Wideband Dual-Channel Linear Multiplier/Dividerapplications. Scaling is accurately determined
by a band-gap voltage reference and all critical pa ..
AD539JN ,Wideband Dual-Channel Linear Multiplier/Dividercharacteristics up
to video frequencies and a 3dB bandwidth of over 60MHz are
provided. Although ..
AD539KD ,Wideband Dual-Channel Linear Multiplier/DividerSpecifications subject to change without notice. boldface are tested on all production units.
-2- ..
AD539KN ,Wideband Dual-Channel Linear Multiplier/DividerAPPLICATIONS
Precise High Bandwidth AGC and VCA Systems
Voltage-Controlled Filters
Video-Sig ..
AD539S ,Wideband Dual-Channel Linear Multiplier/Dividerapplications where differential
phase is critical a reduced input range of t 1 volt is recommended ..
AD539SD ,Wideband Dual-Channel Linear Multiplier/Dividerapplications, with low crosstalk between channels. Voltage-
controlled filters and oscillators usi ..
AD9501JP ,Digitally Programmable Delay GeneratorSPECIFICATIONS1ABSOLUTE MAXIMUM RATINGS Operating Temperature RangePositive Supply Voltage . . . . ..
AD9502AM ,Hybrid RS-170 Video Digitizercharacteristics
increase the tlexibility of the device by making it usable ovcr a
wide range of i ..
AD9502BM ,Hybrid RS-170 Video DigitizerFEATURES
a-Bit Gray Scale Rnsolntion
Screen Resolution m M2 x 512
Phttt-Lottkad Pixel Clock
..
AD9502CM ,Hybrid RS-170 Video DigitizerGENERAL DESCRIPTION
The Analog Devices' AD9502 is a video digitizer which converts
RS-170, NTSC ..
AD9512BCPZ-REEL7 , 1.2 GHz Clock Distribution IC, 1.6 GHz Inputs, Dividers, Delay Adjust, Five Outputs
AD9512BCPZ-REEL7 , 1.2 GHz Clock Distribution IC, 1.6 GHz Inputs, Dividers, Delay Adjust, Five Outputs
AD539JD-AD539JN-AD539KD-AD539KN -AD539S-AD539SD
Wideband Dual-Channel Linear Multiplier/Divider
ANALOG
DEVICES
Wideband Dual-Channel
Linear Multiplier/Divider
FEATURES
Two Quadrant Multiplication/Division
Two Independent Signal Channels
Signal Bandwidth of 60MHa (low)
Linear Control Channel Bandwidth of 5MHz
Low Distortion (to 0.01%)
Fully-Calibrated, Monolithic Circuit
APPLICATIONS
Precise High Bandwidth AGC and VCA Systems
Voltage-Controlled Filters
Video-Signal Processing
High-Speed Analog Division
Automatic Signal-Leveling
Square-Law Gain/Loss Control
PRODUCT DESCRIPTION
The AD539 is a low-distortion analog multiplier having two
identical signal channels (Y1 and Y2), with a common X-input
providing linear control of gain. Excellent ac characteristics up
to video frequencies and a 3dB bandwidth of over 60MHz are
provided. Although intended primarily for applications where
speed is important the circuit exhibits good static accuracy in
"computational" applications. Scaling is accurately determined
by a band-gap voltage reference and all critical parameters are
laser-trimmed during manufacture.
The full bandwidth can be realized over most of the gain range
using the AD539 with simple resistive loads of up to 1000.
Output voltage is restricted to a few hundred millivolts under
these conditions. Using external op amps such as the AD5539 in
conjunction with the on-chip scaling resistors, accurate multipli-
cation can be achieved, with bandwidths typically as high as
50MHz.
The two channels provide flexibility. In single-channel applications
they may be used in parallel, to double the output current, or in
series, to achieve a square-law gain function with a control
range of over 100dB, or differentially, to reduce distortion.
Alternatively, they may be used independently, as in audio
stereo applications, with low crosstalk between channels. Voltage-
controlled filters and oscillators using the "state-variable" approach
are easily designed, taking advantage of the dual channels and
common control. The AD539 can also be configured as a divider
with signal bandwidths up to lSMHz.
Power consumption is only 135mW using the recommended
t 5V supplies. The AD539 is available in three versions: the
"J'' and "K" grades are specified for 0 to +70°C operation and
"S'' grade is guaranteed over the extended range of - 55°C to
+ 125°C. The J and K grades are available in either a hermetic
ceramic DIP (D) or a low cost plastic DIP (N), while the S
REV. A
Information furnished by Analog Devices is believed to be accu rate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
PIN CONFIGURATION
" OUTPUT
Vx (common Li]
HF COMP
MULTIPLY
v" (CHAN1 INPUT) Ei-ey
COMMON
-Vs E E
vniannzmpunE.® 11 (i'lAli,
MULTIPLV
INPUT 6k
COMMON 7 -NNN- E 22
OUTPUT " El W2
COMMON
grade is available in ceramic DIP (D) or LCC (E). J-grade chips
are also available. The S grade is now available in MIL-STD-883
and Standard Military Drawing (DESC) N umber 5962-
8980901EA versions.
DUAL SIGNAL CHANNELS
The signal voltage inputs, V“ and Vw, have nominal full-scale
(FS) values of t 2V with a peak range to 1 4.2V (using a negative
supply of 7. 5V or greater). For video applications where differential
phase is critical a reduced input range of t 1 volt is recommended,
resulting in a phase variation of typically t0.20 at 3.579MHz
for full gain. The input impedance is typically 400kn shunted
by 3pF. Signal channel distortion is typically well under 0.1% at
lOkHz and can be reduced to 0.01% by using the channels
differentially.
_ COMMON CONTROL CHANNEL
The control channel accepts positive inputs, Vx, from 0 to + 3V
FS, t 3.3V peak. The input resistance is 500n. An external,
grounded capacitor determines the small-signal bandwidth and
recovery time of the control amplifier; the minimum value of
3nF allows a bandwidth at mid-gain of about 5MHz. Larger
compensation capacitors slow the control channel but improve
the high-frequency performance of the signal channels.
FLEXIBLE SCALING
Using either one or two external op amps in conjunction with
the on-chip 6ki2 scaling resistors, the output currents (nominally
t lmA FS, t2.25mA peak) can be converted to voltages with
accurate transfer functions of Vw = -VxVv/2, Vw = -VxVy
or Vw = - ZVXVY (where inputs Vx and Irv and output Vw
are expressed in volts), with corresponding full-scale outputs of
t 3V, t 6V and i 12V. Alternatively, low-impedance grounded
loads can be used to achieve the full signal bandwidth of 60MHz,
in which mode the scaling is less accurate.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, USA.
Tel: 617/329-4700 Fax: 617/326-8703 wa: 710/394-6577
Telex: 924491 Cable: ANALOG NORWOODMASS
A0539 - SPEC I Fl Mil illG i@h= 25°C. h-- : w, unless otherwise speeifted)
A0539] AD539K AD539S
Parameter Conditions Min Typ Max Min Typ Max Min Typ Max Units
SIGNAL-CHANNEL DYNAMICS
Minimal Ctsnfiguration Reference Figure 6a
Bandwidth, - 3dB Rt, = SOILCC - 0.01pF 30 60 30 60 30 60 MHz
Maximum Output + 0. lV
Feedthrough, f<1MHz Vx = 0, Vrac = 1.5V rms - 75 - 75 - 75 dBm
f=20MHz -55 -55 -55 dBm
Differential Phase Linearity
-1V-2VGroup Delay Vx = + 3V, Vyac = IV ms, f= IMHz 4 4 4 ns
Standard Dual-Channel Multiplier Reference Figure 2
Maximum Output vx= +3V,Vyac= 1.5Vrms 4.5 4.5 4.5 v
Feedthrough,f<100kHz Vx---0,Vvac---l.5Vrrns 1 1 1 mV rms
Chosstalk(CH1toCH2) Vy1=1Vrms, 1% =0
Vx= +3V,f<:l00kHz -40 -40 -40 dB
RTO Noise, 10Hz to 1MHz Vx = + 1.5V, Ve = 0, Figure 2 200 200 200 nV/VIE
THD + Noise, Vx = +1V, {:10kHz, Vvac -lVrms 0.02 0.02 0.02 %
Vy = + 3V f=101Wide Band Two-Channel Multiplier Figure 2
Bandwidth, - 3dB (LH0032) +0.1VMaximum Output Vx = + 3V Vvac = 1.SV rms, f= 3MHz 4.5 4.5 4.5 V rms
Feedthrough Vx = 0V Vvac = 1.0V ms, f= 3MHz 14 14 14 mV mu
Wide Band Single Channel VCA
(AD5539) Reference Figure 8
Bandwidth, - 3dB + 0.1VMaximum Output 750 Load t 1 t 1 ' 1 V
Feedthroygh Vx = 7 0.01V, f-- SMHz - 54 54 - 54 dB
CONTROL CHANNEL DYNAMICS
Bandwidth, 7 3dB ct = 3000pF, dec = + 1.5V,
Vxac = 100mV ms 5 5 5 MHz
SIGNAL INPUTS, Vy1& Irv,
Nominal Full-Scale Input Y: 2 - 2 t 2 V
Operational Range, Degraded Performance - Vss7V + 4.2 ' 4.2 t 4.2 V
Input Resistance 400 400 kn
Bias Current 10 30 10 20 10 30 "
Offset Voltage Vx = + 3v, vv = o 5 20 5 10 20 mV
(Tm, to Tm) 10 5 15 35 mV
Power Supply Sensitivity Vx _ + 3V, Vy = 0 2 2 2 mVN
CONTROL INPUT, Vx
Nominal Full-Scale Input + 3.0 + 3.0 + 3.0 V
Operational Range, Degraded Performance + 3.2 + 3.2 + 3.2 V
Input Resistance' 500 500 500 n
Offset Voltage l 4 1 2 l 4 mV
(Tm to Tm) 3 2 2 5 mV
Power Supply Sensitivity 30 30 30 pV/V
Gain (Figure 2)
Absolute Gain Error Vx = , 0. 1V to + 3.0V and 0.2 0.4 0.1 0.2 0.2 0.4 dB
(Tm... tonn) v,-- t2V 0.3 0.15 0.25 0.5 dB
CURRENT OUTPUT'
Full-Scale Output Current Vx = + 3V, Vy = 1 2V t 1 , 1 t 1 mA
PeakoutputCurrent Vx---- +3.3V,Vy= t5V,Vs= :7.SV +2 12.8 t2 t2.8 t2 $2.8 mA
Output OffsetCurrent Vx =0, Vy = 0 0.2 1.5 0.2 1.5 0.2 1.5 WA
Output Offset Voltaga2 Figure 2, Vx 2 0, Vy = 0 3 10 3 10 3 10 mV
Output Resistance' 1.2 1.2 1.2 kn
Scaling Resistors
cm Z1,W1 toCHl 6 6 6 kn
CH2 Z2, W2 to CH2 6 6 6 kn
VOLTAGE 0UTPUTS,Vw, 5:sz2 (Figure 2)
Multiplier Transfer Function,
EitherChannel Vw- - Vx-Vy/Vu Vw--- -VxNv/Vu Vw= -vxGrv/Vu
Multiplier Scaling Voltage, Vu 0.98 1.0 1.02 0.99 1.0 1.01 0.98 1.0 1.02 V
Accuracy 0. 5 2 0. 5 l 0.5 2 %
(TM to Tm) 1 0.5 1.0 , %
Power Supply Sensitivity 0.04 0.04 0.04 %/V
Total Multiplication Errors Vx< = + 3V, 7 2VTis,, to TU, 2 l 2 4 %
Control Feedthrough Vx _ 0to 1 3V, Ve =0 25 60 15 30 15 60 mV
Tm to Tu, 30 15 60 120 mV
TEMPERATURE RANGE
Rated Performance 0 + 70 0 + 70 - 55 + 125 'tl
POWER SUPPLIES
OperationalRange 14.5 :15 +4.5 :15 +4.5 :15 V
Current Consumption
+Vs 8.5 I0.2 8.5 10.2 8.5 10.2 mA
-Vs 18.5 22.2 18.5 22.2 18.5 22.2 M
'Resisunce value and absolute current outputs subject to 20% tolerance. Stmcificatiuns shown in boldface are tested on all production units " final electri-
ISpec assumes the external op amp is trimmed for negligible input offset. cal test. Results from (hos: tests are used to calculate outgoing quality levels. All
'Inchuies all em“, min and max specifications are guaranteed, although only those shown in
Specifications subiecr to change without notice. boldface ttre tested on all praiuction units.
-2- R EV. A
ORDERING GUIDE
Temperature Package Package
Model Range Description Option _
AD539JN 0°C to + 70°C Plastic DIP N-16
AD539KN 0°C to + 70°C Plastic DIP N-16
AD539JD 0°C to + 70°C Side Brazed DIP D-l6
AD539KD 0°C to + 70°C Side Brazed DIP D-16
AD539J Chip 0°C to + 70°C Chip
AD539SD - 55°C to + 125°C Side Brazed DIP D-16
AD539SD/883B - 55°C to + 125°C Side Brazed DIP D-l6
5962-89g0901EAI - 55°C to + 125°C Side Brazed DIP D-16
AD539SE/883B - 55°C to + 125°C LCC E-20A
lThe standard military drawing version of the AD539 (5962-898090IEA) is now available.
Ve, Vw, =
_VX'VYI
Vx EXTERNAL
OP AMPS
Y2 - Vx'sz
AD539 Functional Block Diagram
CIRCUIT DESCRIPTION
Figure l is a simplified schematic of the AD539. Q1-Q6 are
large-geometry transistors designed for low distortion and low
noise. Emitter-area scaling further reduces distortion: Q1 is 3
times larger than Q2; Q4, Q5 are each 3 times larger than Q3,
Q6, and these transistors are twice as large as Q1, Q2. A stable
reference current IREF = 1.375mA is produced by a band-gap
reference circuit and applied to the common emitter node of a
controlled-cascade formed by Q1 and Q2. When Vx = 0, all of
[REF flows in Q1, due to the action of the high-gain control
amplifier which lowers the voltage on the base of Q2. As Vx is
raised the fraction of IREF flowing in Q2 is forced to balance the
control current, Vx/2.5k. At the full-scale value of Vx (+ 3V)
this fraction is 0.873. Since the bases of Q1, Q4 and Q5 are at
ground potential and the bases of Q2, Q3 and Q6 are commoned,
all three controlled-cascades divide the current applied to their
emitter nodes in the same proportion. The control loop is stabilized
by the external capacitor, Cc.
CONTROL
INPUT t
o TO oav FS
2 " coumol l me L250!
Ammslsn a
_ cm " m Wt m E H2
L2,... ourpur ' El Ls I ' OUTPUT
rs " "
ti-sl-s/HEI'' "EFF-te
a OmPUT
COMMON
= th, mm
W = T 3nF min
I.375'nA l -
' cm em
Vs BANU-t9AP INPUT n nan
REFERENCE 4 HV rs :zv FS
V GENERATOR
INPUT COMMON
Figure I. Simplified Schematic of A0539 Multiplier
REV. A
The signal voltages VYl and Vve (generically referred to as VY)
are first converted to currents by voltage-to-current converters
with a gm of 57hunhos; thus, the full-scale input of t 2V becomes
a current of i 1.15mA, which is superimposed on a bias of
2.75mA, and applied to the common emitter node of controlled
cascade Q3-Q4 or Q5-Q6. As just explained, the proportion of
this current steered to the output node is linearly dependent on
Vx. Thus for full-scale Vx and Vy inputs, a signal of t lmA
(0.873 X t 1.15mA) and a bias component of 2.4mA (0.873 X
2.75mA) appear at the output. The bias component absorbed by
the 1.25k resistors also connected to Vx, and the resulting signal
current can be applied to an external load resistor (in which case
scaling is not accurate) or can be forced into either or both of
the 6kn feedback resistors (to the Z and W nodes) by an external
op amp. In the latter case, scaling accuracy is guaranteed.
GENERAL RECOMMENDATIONS
The AD539 is a high speed circuit and requires considerable
care to achieve its full performance potential. A high-quality
ground plane should be used with the device either soldered
directly into the board or mounted in a low-profile socket. In
the figures used here an open triangle denotes a direct, short
connection to this ground plane; pins 12 and 13 are especially
prone to unwanted signal pick-up. Power supply decoupling
capacitors of 0. luF to 1p.F should be connected from pins 4
and 5 to the ground plane. In applications using external high-speed
op amps, separate supply decoupling should be used. It is good
practice to insert small (10fl) resistors between the primary
supply and the decoupling capacitor.'
The control amplifier compensation capacitor, Cc, should likewise
have short leads to ground and a minimum value of 3nF. Unless
maximum control bandwidth is essential it is advisable to use a
larger value of 0.01p.F to 0.1WF to improve the signal channel
phase response, high-frequency crosstalk and high-frequency
distortion. The control bandwidth is inversely proportional to
this capacitance, typically 2MHz for Cc = 0.01WF, Vx = 1.7V.
The bandwidth and pulse response of the control channel can
be improved by using a feedforward capacitor of 5% to 20% the
value of CC between pins 1 and 2. Optimum transient response
will result when the rise/fall time of Vx are commensurate with
the control-channel response time.
Vx should not exceed the specified range of 0 to + 3V. The ac
gain is zero for Fret but there remains a feedforward path (see
Figure I) causing control feedthrough. Recovery time from
negative values of Vx can be improved by adding a small-signal
Schottky diode with its cathode connected to pin 2 and its anode
grounded. This constrains the voltage swing on CC. Above Vx
= + 3.2V, the ac gain limits at its maximum value, but any
overdrive appears as control feedthrough at the output.
The power supplies to the AD539 can be as low as 14.5V and
as high as i 16.5V. The maximum allowable range of the signal
inputs, Irv, is approximately 0.5V above + Vs; the minimum
value is 2.5V above -Vs. To accommodate the peak specified
inputs of t4.2V the supplies should be nominally + 5V and
_ 7.5V. While there is no performance advantage in raising
supplies above these values, it may often be convenient to use
the same supplies as for the op amps. The AD539 can tolerate
the excess voltage with only a slight effect on dc accuracy but
dissipation at : 16.5V can be as high as 535mW and some form
of heat-sink is essential in the interests of reliability.
TRANSFER FUNCTION
In using any analog multiplier or divider careful attention must
be paid to the matter of scaling, particularly in computational
applications. To be dimensionally consistent a scaling voltage must
appear in the transfer function, which, for each channel of the
AD539 in the standard multiplier configuration (Figure 2) is
Vw = "vaY/V U
where the inputs Vx and Vy, the output Vw and the scaling
voltage VU are expressed in a consistent unit, usually volts. In
this case, Vu is fixed by the design to be 1V and it is often
acceptable in the interest of simplification to use the less rigorous
expression
Vw = -VxVy
where it is understood that all signals must be expressed in volts,
that is, they are rendered dimensionless by division by (IV).
The accuracy specifications for Irv allow the use of either of the
two feedback resistors supplied with each channel, since these
are very closely matched, or they may be used in parallel to
halve the gain (double the effective scaling voltage), when
Vw = -VxVv/2.
When an external load resistor, RL, is used the scaling is no
longer exact since the internal thin-film resistors, while trimmed
to high ratiometric accuracy, have an absolute tolerance of 20%.
However, the nominal transfer function is
Vw = -VxVv/Vv'
where the effective scaling voltage, Vc' can be calculated for
each channel using the formula Vu' = T, (SRL + 6.25)/RL,
where RL is expressed in kilohms. For example, when RL =
1000, Vc' = 67.5V. Table II provides more detailed data for
the case where both channels are used in parallel. The AD539
can also be used with no external load (output pin 11 or 14
open-circuit), when Vu' is quite accurately 5V.
BASIC MULTIPLIER CONNECTIONS
Figure 2 shows the connections for the standard two-channel
multiplier, using op amps to provide useful output power and
the AD539 feedback resistors to achieve accurate scaling. The
transfer function for each channel is
Vw = -VxVv
where inputs and outputs are expressed in volts (see TRANSFER
FUNCTION). At the nominal full-scale inputs of Vx = + 3V,
Vy = 1 2V the full-scale outputs are 16V. Depending on the
choice of op amp, their supply voltages usually need to be about
2V more than the peak output. Thus, supplies of at least t 8V
are required; the AD539 can share these supplies. Higher outputs
are possible if Vx and Vy are driven to their peak values of
CONTROL W1
f‘l F-HE HF COMP 21
cm A9539 CH1
INPUT OUTPUT
+ Vs BASE
- Vs COMMON
CH2 CH2
INPUT OUTPUT
INPUT "
COMMON
OUTPUT
COMMON W2
ALL DECOUPLING CAPACITORS ARE 0.47pF CERAMIC.
Figure 2. Standard Dual-Channel Multiplier
+ 3.2V and : 4.2V respectively, when the peak output is , 13.4V.
This requires operating the op amps at supplies of t 15V. Under
these conditions it is advisable to reduce the supplies to the
AD539 to t7.5V to limit its power dissipation; however, with
some form of heat sinking it is permissible to operate the AD539
directly from t 15V supplies.
Viewed as a voltage-controlled amplifier, the decibel gain is
simply
G = 20log Vx
where Vx is expressed in volts. This results in a gain of 10dB at
Vx = +3.162V, OdB at Vx = +1V, --20dB at Vx = +0.1V,
and so on. In many ac applications the output offset voltage (for
Vx = 0 or VY = O) will not be of major concern; however, it
can be eliminated using the offset mulling method recommended
for the particular op amp, with Vx = Irv = 0.
At small values of Vx the offset voltage of the control channel
will degrade the gain/loss accuracy. For example, a t lmV
offset uncertainty will cause the nominal 40dB attenuation at Vx
= + 0.01V to range from 39.2dB to 40.9dB. Figure 3a shows
the maximum gain error boundaries based on the guaranteed
control-channel offset voltages of t2rnV for the AD539K and
t4mV for the AD539]. These curves include all scaling errors
and apply to all configurations using the internal feedback resistors
(W1 and W2; alternatively, ZI and Z2).
ADSIISJ. s
GAINILOSS ERROR - d5
+ Fo, + IM + , + 10
CONTROL VOLTAGE - Vx
Figure 3a. Maximum AC Gain Error Boundaries
Distortion is a function of the signal input level (VY) and the
control input (Vx). It is also a function of frequency, although
in practice the op amp will generate most of the distortion at
frequencies above 100kHz. Figure 3b shows typical results at
f = lOkHz as a function of Vx with VY = 0.5 and 1.5V rms.
UL-.-----'")
l=10kHz
TOIAL HARMONIC DISTORTION - %
CONTROL VOLTAGE - V
Figure 3b. Total Harmonic Distortion vs. Control Voltage
REV. A
In some cases it may be desirable to alter the scaling. This can
be achieved in several ways. One option is to use both the Z
and W feedback resistors (see Figure l) in parallel, in which
case Vw = - VxVv/2. This may be preferable where the output
swing must be held at t 3V FS (16.75V pk), for example, to
allow the use of reduced supply voltages for the op amps. Alter-
natively, the gain can be doubled by connecting both channels t
in parallel and using only a single feedback resistor, in which
case Vw = - ZVXYY and the full-scale output is t 12V. Another
option is to insert a resistor in series with the control-channel
input, permitting the use of a large (for example, 0 to + 10V)
control voltage. A disadvantage of this scheme is the need to
adjust this resistor to accommodate the tolerance of the nominal
5000 input resistance at pin I. The signal channel inputs can
also be resistively attenuated to permit operation at higher values
of Vy, in which case it may often be possible to partially compensate
for the response roll-off of the op amp by adding a capacitor
across the upper arm of this attenuator.
Signal-Channel ac and Transient Response
The HF response is dependent almost entirely on the op amp.
Note that the "noise gain" for the op amp in Figure 2 is determined
by the value of the feedback resistor (6kn) and the 1.25kf1
control-bias resistors (Figure I), Op amps with provision for
external frequency compensation (such as the AD301 and AD518)
should be compensated for a closed-loop gain of 6.
The layout of the circuit components is very important if low
feedthrough and flat response at low values of Vx is to be main-
tained (see GENERAL RECOMMENDATIONS).
For wide-bandwidth applications requiring an output voltage
swing greater than 1 IV, the LH0032 hybrid op-amp is recom-
mended. Figure 4a shows the HF response of the circuit of
Vx = 3.162V
Vx = 1.00V
vx=o.a1sv
dB -20
Vx--0032mV
Vx = 0.01V
FEEDTHROUGH
-50 v, -- -0.01V
100k 1M 10M
FREQUENCY - Hz
Figure 4a. Multiplier HF Response Using LH0032 Op
Figure 2 using this amplifier with v, = IV rms and other
conditions as shown in Table I. c, was adjusted for ldB peaking
at Vx = +1V; the - 3dB bandwidth exceeds 25MHz. The
effect of signal feedthrough on the response becomes apparent
at Vx = +0.01V. The minimum feedthrough results when Vx
is taken slightly negative to ensure that the residual control-channel
offset is exceeded and the dc gain is reliably zero. Measurements
show that the feedthrough can be held to -90dB relative to full
output at low frequencies and to - 60dB up to 20MHz with
careful board layout. The corresponding pulse response is showri
in Figure 4b for a signal input of Vy of 1 1V and two values of
Vx (+ 3V and +0.1V).
Vx = +3V
Figure 4b. Multiplier Pulse Response Using LH0032 Op
AD711l AD55392 LH0032l
Op Amp Supply Voltages 1 15V t 9V 1 10V
Op Amp Compensation Capacitor None None l-SpF
Feedback Capacitor, Cr, None 0.25-1.5pF 1-4pF
- 3dB Bandwidth, Vx = + 1V 900kHz SOMHZ 25MHz
Load Capacitance < 1 nF <10pF <100pF
HF Feedthrough,
Vx = -0.0IV,f = 5MHz N/A - S4dB - 70dB
rms Output Noise,
Vx = + lV,BWlOHz-10kHz 50w 40w 30w
Vx = + 1V, BW 10Hz-5MHz 120p.V 620p.V 500wV
In all cases, 0.47p.F ceramic supply-decoupling capacitors were used at each IC
pin, the AD539 supplies were t 5V and the control-compensation capacitor Cc
was 3nF.
'For the circuit of Figure 2.
Tor the circuit of Figure 8.
Table l. Summary of Operating Conditions and Perform-
ance for the A0539 When Used with Various External
Op-Amp Output Amplifiers
Minimal Wide-Band Configuratipns
The maximum bandwidth can be achieved using the AD539
with simple resistive loads to convert the output currents to
voltages. These currents (nominally It: lmA FS, t 2.25mA pk,
Table II. Summary of Performance for Minimal Configuration
Load Resistance son 750. 1000 1500. 6009. 0/C
FS Output Voltage t 92.6mV t l34mV t 172mV t 242mV t 612mV t 1V
65.5mV rms 94.7mV rms 122mV rms l7lmV rms 433mV rms *
FS Output- 0.086mW 0.12mW 0.15mW 0.195mW 0.312mW -
Power in Load 1 10.5dBm - 9.2dBm - 8.3dBm - 7.1dBm - 5.05dBm -
Pk Output Voltage 1 210mV 1 300mV 1 388mV 1 544mV t 1V t 1V
l48mV rms 212mV rms 274mV rms 385mV rms * *
Pk Output- 0.44mW 0.6mW 0.75mW lmW 11V 11V
Power in Load - 7dBm - 4.4dBm - 2.5dBm OdBm A *
Effective Scaling 67.5V 46. 7V 36.3V 25.8V 10.2V 5V
Voltage, VU’
*Peak negative voltage swing limited by output compliance.
R EV. A -5-
AIIA I An l‘l'lllnl'l‘
