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LM1837NNSC N/a2000avai18 V, 9 mA, low noise preamplifier for autoreversing tape playback system


LM1837N ,18 V, 9 mA, low noise preamplifier for autoreversing tape playback systemFeatures I Programmable turn-on delay I Transient-free power-uP-no pops l Transient-free mut ..
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LM1837N
18 V, 9 mA, low noise preamplifier for autoreversing tape playback system
LM1837
National '
I semiconductor
LM1837 Low Noise Preamplifier for
Autoreversing Tape Playback Systems
General Description
The LM1837 is a dual autoreversing high gain tape pream-
plifier for applications requiring optimum noise performance.
It has forward (left, right) and reverse (left. right) inputs
which are selectable through a high impedance logic pin. It
is an ideal choice for a tape playback amplifier when a com-
bination of low noise, autoreversing, good power supply re-
jection, and no power-up transients are desired. The appli-
cation also provides transient-free muting with a single polo
grounding switch.
Features
a Programmable tum-on delay
I: Transient-free power-utr-no pops
n Transient-tree muting
Low noiso-4h6 WV CCIR/ARM in a DIN circuit refer-
enced to 1 kHz
Low voltage battery operation -4V
Wide gain bandwidth due to broadband two-amplifier
approach-N dB g 20 kHz
High power supply "isction-95 dB
Low distortion-AMP/o
Fast slew rate-AN/ws
Short circuit protection
Internal diodes for diode switching applications
Low cost external parts
Excellent low frequency response
Prevents "click'' from being recorded onto the tape dur-
ing power supply cycling in tape playback applications
High impedance logic pin for forward/reverse switching
Vcc=12 Vac " um
1.2" 5%
swam" Pi81
WIRE RIGHT
" (an I i)?: . oumn
mm mmmn "N,, as
INPUT 10k
l] ' iysr
RIGHT REVERSE I I
INPUT -- LOGIC
mgr I I
I I x25
LEFTFORWABD l I o
mm I l 's, m
4711 pF
(Wk Len
t l M " ' OUTPUT
. , = -cit:s---o-tyr- ---o
LEF‘I'REVEHSE I I
MOT m pF I I
tt 10 13 (11.3) a :15” 'a
JI "rf M 0.0022 pr m
Mi-- AY tNt
. LOGIC '
Numbers m parentheses are for FthtBhft0sll.W . 38
Smalloutline package. REVERSEzZJV IN TWIF tlf?- k" r,t' tPtttNAL MUTE
'Not bonded out in Small Outline - " 1
package. -
- TL/H/7902-1
FIGURE 1. Autoreversing Tape Playback Appllcatlon
Absolute Maximum Ratings
" Military/Aerospace ttpttelfied devices are required,
please contact the Natlonal Semlconductor Sales
OfflttefDlatrlbutora for avallablllty and specifications.
Supply Voltage 18V
Voltage on Pins 1 and 18 18V
Package Dissipation (Note 1) 1390 mW
Storage Temperature - 65''C to + 150°C
Soldering Information
Dual-ln-Line Package
Soldering (10 seconds) 260°C
Small Outline Package
Vapor Phase (60 seconds) 215''C
Infrared (15 seconds) 220°C
See AN-450 "Surface Mounting Methods and Their Effect
on Product Reliability" for other methods of soldering sur-
Operating Temperature
Minimum Voltage on Any Pin
Electrical Characteristics (TA = 25''C, Vcc = 12V, see TestCircuit, Figural?)
UC to + 7ty'C
-0.1 VDC
face mount devices.
Parameter Conditions Min Typ Max Unlts
Operating Supply Voltage Range R5 Removed from Circuit for
. 4 18 V
Low Voltage Operation
Supply Current Vcc = 12V 9 15 mA
Total Harmonic Distortion f = 1 kHz, VIN = 0.3 mV, 0 03 o/
Pins 2 and 17, Figure 2 . °
THD + Noise (Note 2) f = 1 kHz, VOUT = IV, q
Pins 2 and 17, Figure 2 0.1 0 th25 A
Power Supply Rejection Input Ref. f = 1 kHz, 1 Vrms 80 95 dB
Channel Separation (Note 3) f = 1 kHz. Output = 1 Vrms, dB
Left to Right Output to Output 40 60 dB
Forward to Reverse 40 60 dB
SignaI-to-Noise (Note 4) Unweighted 32 Hz-12.74 kHz (Note 2) 58 dB
CCIR/ARM (Note 5) 62 dB
A Weighted 64 dB
CCIR, Peak (Note 6) 52 dB
Noise Output Voltage CCIR/ARM (Note 5) 120 200 p.V
Input Amplifers
Input Bias Current 0.5 2.0 p.A
Input Impedance f = 1 kHz 150 kn
AC Gain 27 28 29 dB
AC Gain imbalance i 0.15 i 0.5 as
DC Output Voltage 2.1 2.5 2.9 V
DC Output Voltage Mismatch Pins 5 and 14 -200 i 30 200 mV
Output Source Current Pins 5 and 14 2 10 mA
Output Sink Current Pins 5 and 14 300 600 p.A
Logic Level
Forward 0.5 V
Reverse 2.2 V
Logic Pin Current 6 p.A
DC Voltage Change at Change Logic State - 100 + 20 100 mV
Pins 5 and 14
LM1837
Electrical Characteristics (TA = 25"C, vcc = 12V, see TestCircuit, Figure21 (Continued)
Parameter Cttrtdltittntt Mln Typ Max Units
Output Amplifiers
Closed Loop Gain Stable Operation 5 WV
Open Loop Voltage Gain DC 100 dB
Gain Bandwidth Product 5 MHz
Slew Rate 6 V/ps
Input Offset Voltage 2 5 mV
Input Offset Current 20 100 nA
Input Bias Current 250 500 nA
Output Source Current Pin 2 or 17 2 10 mA
Output Sink Current Pin 2 or 17 400 900 p.A
Outut Voltage Swing Pin 2 or 17 1 1 Vp-p
Output Diode Leakage Voltage on Pins 1 and 18 = 18V 0 10 p.A
Note 1: For operation In ambient temperatures above 25'C,1he device must be derated based on a 160% maximum lunctlon temperature and a thermal resistance
ot QO'CIW junctlon to emblem (DuaI-ln-Line). Small Outline Thermal Reslatance is 100‘C/W.
Note 2: Measured with an average responding voltmeter using the filter drcult in Figure 4. Thls simple tlltttr lit approximately equivalent to a "brick wall" fltter wlth a
passbend of 20 Hz to 20 kHz (see Application Hints). For 1 kHz THD the 400 Hz high pass filter on the distortion enelymr Ia used.
Note & Channel separation can be measured by applying the Input slgnal through tranaformtrrg to simulate a floating source (see Appllcatlon Hints). Care must be
taken to ahletd the coils from extraneous slgnals. Actual production test technlquee at National simulate th1s floatlng source with a more complex op amp circuit
Note 4: The numbers are referred to an output level of 160 mV at pins 2 and 17 using the circuit of FlgureZ. Thls corresponds to an Input level of 0.3 mes at 333
Note 5: Measured with an average responding voltmeter using the Dolby Iab's standard CCIR filter having a unity gain reference at 2 kHz.
Note a.. Measured using the Rhode-Schwarz peophometer. model UPGR.
Typical Performance Characteristics
Input Amplifier THD
vslnput Level
F " ttmnawruria 0111.1
S to 1.11111: 3"
g E 10
I " S 0
E t.0 -10
, -esess--"-" -N
0 N w 00 00 100
MrllT0tihtttt)
Spot Noise Voltage
vs Fre uenc
100 q Y
10 100 " IN
msnuaacv 111:)
Turn-On Delay "
Component Values
and Galn
'a'! 300
il 250
i 'iii
it 150 E
II " 0.4 " " 1.0 1.2
YUM?! “NE DELAY (SECMDS)
Input Amp0tsr Galn and
Phase "
’3 8 S 8 °
(mumnsvm
10 100 lk 10k 100K "I 10M1WM
F1tEtiilEiltt (11;)
Spot Nolse Current
0 vs Frequency
10 100 lk IN
11150091011112)
PSRR vs Frequency
IN mm! REFERRED
cmcun " FIGURE t
40 Vac =12V
N 50100200500 " tk 5111011201
msuumcv (H1)
PSI"! (03)
Output Amplifier Open
Loop Gain and Phase
ao vs Frequency
PHASE 'lt
240 E.
10 100 " 101110011 111 101110011
smusucv 1111)
TL/H/7902-2
Total Harmonlc Dlgtttrtlttrt
vs Frequency
F " 01110011 OF FIGURE!
G" " V001=1 htttt
il-', 2.0
'ie, 41110110511
1: 20 112-20 kHz mm!
i 0.05
tt thot
N 100 500 " tttk
mmmcv 1111;
TLfH/7902-3
PSRR " Vcc
60 11170111331120
macun 0F FIGURE 2
10 115 1151101150
i=1 tttz
- as @1 101:
N . l . l
1 6 1110121111118
Vw (V)
TL/H/7902-4
£88 l W'l
LM1837
Typical Performance Characteristics (Continued)
Icc " VCC
lcc (ml)
Connection Diagrams
R5 REMOVED
vec (V)
Right to Left Channel
Separation " Frequency
CHANNEL SEPARATION (dB)
taittah is niiimi 2
"s, ts,
20 50100200 Mole 2k 5k10k20k
rnmumcv (H1)
Input Ampllfler Dc Output
Voltage vs Temperature
(Pins 5, 14)
9mm nc oumn vomss (V)
-S0--N D " MI 75100IN
Small Outline Package
RIGHT OUTPUT - 1 16 - LEFTOUTPUT
R(+)IN- 2 15 -L(o)N
R(-)IN- 3 " --L(-)IN
RIGHT X25 OUT - 4 13 -LUT X25 OUT
BIAS - 5 12 - LOGIC
RIGHT FORWARDINPUT - 6 11 - LEFT FORWARDINPUT
R|GHT REYSRS0INT - 7 10 - LEFT REVERSEINPUT
" - 8 9 - 6ND
TL/H/7902-7
Order Number LM1837M
See NS Package Number M163
TEMPERATURE (oil)
TL/H/7902-6
Forward to Reverse
Channel Separation
" Frequency
tititairr or rlduns t
N 501002005001: " smuzok
ramencvmz)
TL/H/7902-5
Dual-ln-Llne Package
RIGHT moo: OUIPUT 2
mam nuwur A
am m l
Rl-) m A
mm = our .h
BIAS A,
mum FORWARD INPUT 2.
HIGHT REVERSE INPUT 1
3 LEFT moo: ourrur
IT. LEFT ourrur
E u + , IN
3-5 u-; m
It LEFT X25 OUT
3-3 LOGIC
Lt LEFI romann INPUT
3 LEFT amass mm
" J!. 2? sun
TL/H/7902-8
Top View
Order Number LM1837N
See NS Package Number N18A
External Components (Figure t)
Component Normal Range of Value and Function
R2, R3
R4, C1
2 kn-40 kn, 0.1 psF-10 p.F (low leakage)
Set turn-on delay and second amplifior's low
frequency pole. Leakage current in C2 results
in DC offset between the amplifitsr's inputs
and therefore this current should be kept low.
R1 is set equal to R2 such that any input off-
set voltage due to bias current is effectively
cancelled. An input offset voltage is generat-
ed by the input offset current multiplied by
the value of these resistors.
2 kn-40 kn, 500 kn-10 Mn
Sets the DC and low frequency gain of the
output amplifier. The total input offset voltage
will also be multiplied by the DC gain of this
amplifier. It is therefore essential to keep the
input offset voltage specification in mind
when employing high DC gain in the output
amplifier; i.e., 5 mV M 400 = 2V offset at the
output.
10 kn-200 kn, 0.00047 psF-0.01 pF
Set tape playback equalization characteris-
tics in conjunction with R3 (calculations for
the component values are included in the Ap-
plication Hints section).
Simplified Schematic
Component Normal Range of Value and Function
2 kn-47 kn
Biases the output diode when it is used in DC
switching applications. This resistor can be
excluded if diode switching is not desired.
100 pF-1000 pF
Often used to resonate with tape head in or-
der to compensate for tape piay-back losses
including tape head gap and eddy current.
For a typical cassette tape head, the reso-
nant frequency selected is usually between
13 kHz and 17 kHz.
100 kn-10 Mn
Increases the output DC bias voltage from
the nominal 2.5V value (see Application
Hints).
Optionally used for tape muting. The use of
this resistor can also provide "no-pop" turn-
off if desired (see Application Hints).
_..._._.....—...._.._o
ml L15
i) 3, " t, " l, "
TL/Hf7902-9
£88l. W'I
LM1837
Application Hints
LIA 1331
WWII ,
a 3—01
00TH."
$EtECT
t lee:
ttht " x25 I 2u,
stun REF I
l" u " lil "
LOGIC rutr,'-jb,
Pumas n.5v IN
mass 2 2.2v 1%
TL/H/7902- 1 t)
FIGURE 2. General Test Clrcult
mm mm w- [SEI
N nus 2 on " -o
ii" " o
I um msmmou WI
it vomnstsn
TL/H/7g02-12
FIGURE 4. Simple 32 Hz-12740 Hz Fllter and Meter
20 501mm soon " 5k10l20k
manual“ um
TL/H/7902-11
FIGURE 3. Frequency Response of
Test Circuit
Application Hints (Continued)
DISTORTION MEASUREMENT METHOD
In order to clearly interpret and compare specifications and
measurements for low noise preamplifiers, it is necessary to
understand several basic concepts of noise. An obvious ex-
ample is the measurement of totai harmonic distortion at
very low input signal levels. Distortion analyzers provide out-
puts which allow viewing of the distortion products on an
oscilloscope. The oscilloscope often reveals that the "dis-
tortion" being measured contains I) distortion, 2) noise, and
3) 50 or 60 cycle AG line hum.
Line hum can be detected by using the "line sync" on the
oscilloscope (horizontal sync selector). The triggering of a
constant waveform indicates that AC line pick-up is present.
This is usually the result of electro-magnetic coupling into
the preamplifiers input or improper test equipment ground-
ing, which simply must be eliminated before making further
measurements!
Input coupling problems can usually be corrected by any
one of the following solutions: 1) shielding the source of the
magnetic field (using mu metal or steel), 2) magnetically
shielding the preamplifier, a) physically moving the preem-
plitier far enough away from the magnetic field, or 4) using a
high pass filter (to = 200 Hz-l kHz) at the output of the
preamplifer to prevent any line signal from entering the dis-
tortion analyzer. Ground loop problems can be solved by
rearranging ground connections of the circuit and test
equipment.
Separating noise from distortion products is necessary
when it is desired to find the actual distortion and not the
eignei-to-noiae ratio of an amplifier. The distortion produced
by the LM1837 is predominantly a second harmonic. it is for
this reason that the third and higher order harmonies can be
filtered without resulting in any appreciable error in the mea-
surement. The filter also reduces the amount of noise in the
measured data. Another more tedious technique for mea-
suring THD is to use a wave analyzer. Each harmonic is
measured and then summed in an rms calculation. A typical
curve is plotted for distortion " frequency using this meth.
od. A typical curve is also included using a 20 Hz to 20 kHz
4th order filter.
To specify the distortion of the LM1837 accurately and also
not require unusual or tedious measurements the following
method Is used. The output level is set to 1 Vrms at 1 kHz
(approximately 5 mV at the input), The output is filtered with
the circuit of Figure 4 to limit the bandwidth of the noise and
measured with a standard distortion analyzer. The analyzer
has a filter that is switched in to remove line hum and
ground loop pick-up " well as unrelated low frequency
noise. The resulting measurement is fast and accurate.
SIGNAL-TO-NOISE RATIO
In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques. they appear equal in measure.
ments. This is often the case when comparing integrated
circuit to discrete preamplifier designs. Discrete transistor
preamps often "run out of gain" at high frequencies and
therefore have small bandwidths to noise as indicated in
Figure 5.
GAIN (dB)
20 200 "
FREQUENCY (ii!)
20k 200k "
TL/H/7g02-13
FIGURE 5
Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic
distortion and Improve frequency response. It is this addi-
tional bandwidth that can lead to erroneous aignai-to-noise
measurements if not considered during the measurement
process. in the typical example above, the difference in
bandwidth appears small on a log scale but the factor of 10
in bandwidth (200 kHz to 2 MHz) can result in a 10 dB
theoretical difference in the signal-to-noise ratio (white
noise is proportional to the square root of the bandwidth in a
system).
In comparing audio amplifiers It is necessary to measure the
magnitude ot noise in the audible bandwidth by using a
"weighting" filter.1 A "weighting" filter alters the frequency
response in order to compensate for the average human
eer's sensitivity to certain undesirable frequency spectra.
The weighting filters at the same time provide the bandwidth
limiting as discussed in the previous paragraph.
The 32 Hz to 12740 Hz filter shown in Figure 4 is a simple
two pole, one zero filter, approxlmately equivalent to a
"brick wall" filter of 20 Hz to 20 kHz. This approximation is
absolutely valid if the noise has a flat energy spectrum over
the frequencies Involved. In other words a measurement of
a noise source with constant spectral density through either
of the two filters would result in the same reading. The out-
put frequency response of the two filters is shown in Figure
1.88”“
LM1837
Application Hints (Continued)
E mmom
" NOISE At
20 20k
Fnauusucv (Hz)
"MICK mu" rum
TL/H/7902-14
g ----tWl0WItmt
ii! tl ,CC"s,
12140 211k
32 ammo fit rum:
TL/H/7902-15
FIGURE 6
Typical signal-to-noise figures are listed for several weight-
ing filters which are commonly used in the measurement of
noise. The shape of all weighting filters is similar with the
peak of the curve usually occurring in the 3 kHz-? kHz re-
gion as shown in Figure 7.
AMPLITUDE (dB)
N 200 " 5k 20k
FREQUENEY (ht)
TL/H/7902-16
FIGURE 7
In addition to noise filtering, differing meter types give differ-
ent noise readings. Meter responses include: I) rms read-
ing, 2) average responding, 3) peak reading, and 4) quasi
peak reading. Although theoretical noise analysis is derived
using true rms (root mean square) based calculations, most
actual measurement is taken with ARM (Average Respond-
ing Meter) test equipment.
Unless otherwise noted an average responding meter is
used for all AC measurements in this data sheet.
BASIC CIRCUIT APPROACH
The LM1837 IC incorporates a two stage broadband design
which minimizes noise, attains overall DC stability and pre-
vents audible transients during turn-on.
The first Mags consists of four direct coupled preamplifiers
with internal gain of 25V/V (28 dB). Direct coupling to the
tape head reduces input source impedance and external
component cost by removing the input coupling capacitor. A
typical input coupling capacitor of 1 “F has a reactance of
1.5 kn at 100 Hz. The resulting noise due to the amplitier's
input noise current can dominate the noise voltage at the
output of the playback system. The inputs of the amplifiers
are biased from a common reference voltage that is temper-
ature compensated to produce a quiescent DC voltage of
2.5V at the output of the first stage. The input stage bias
current that flows through the tape head is kept below 2 p.A
in order to prevent any erasure of tape moving past the
head. An added advantage of DC biasing is the prevention
of large current transients during the charging of coupling
capacitors at turn-on and turn-oft. The outputs of the for-
ward and reverse preamplifier are fed to the common output
op amp through a logic controlled switch.
The second stage provides additional gain and proper
equalization while preventing audible turn-on transients or
"pops". The output (pin 2) is kept low until C2 charges
through RI. When the voltage on C2 gets close to the DC
voltage on pin 5, the output rises exponentially to its final
DC value. The result is a transient-free turn-on characteris-
Internal diodes are provided to facilitate electronic diode
switching, popular in automotive applications.
The General Test Circuit illustrates the topography of the
system. The components determining the overall frequency
response are external due to the extreme sensitivity when
matching a DIN equalization curve.
MUTE CIRCUIT AND LOGIC
The LM1837 can be muted with the addition of two resistors
and a grounding switch, as shown in Figure 1. When the
circuit is not muted the additional resistors have no effect on
the AC performance. They do have an effect on the DC Q
point however.
The difference in the DC output voltages of the input ampliti-
ers is applied across the mute resistors (R7) and the posi-
tive input resistors (R1). This results in an additional offset
at the input of the output amplifiers. To keep this offset to a
minimum R7 should be as large as possible to achieve ef-
fective muting. Unmute voltage is the peak signal the pre-
amplifier can swing without turning on the output amplifier
under mute conditions:
R5//R3
Unmute - R7 1
R2 + R5//R3 R1 + R7
voltage = VP‘N 5, 14 [
Application Hints (Continued)
For example: The circuit in Figure 1 has 2.5V DC at pins 5
and 14, so:
Unmute voltage ---
1.2M//1.5M - 270k
10k + 1.2M//1.5M 10k + 270K
It may be necessary to slow the transition of the logic pin it
the mute circuit is not used. The forward and reverse pre-
amplifier output DC voltages can differ by , 100 mV. This
rapid DC charge is gained up by the output amplifier and
appears as a pop. The circuit of Figure tt will slow the DC
transition.
2.5V [ 1 = 52.3 mV
V*=12V
TO FIN 13
TL/H/7902-17
FIGURE 8. Circuit to Slow Logic
DESIGN EQUATIONS
The overall gain of the circuit is given by:
-R4R3 (s + Wm)
R2(R3 + M) ( 1 ) (1)
(R3 + R4)C1
Standard cassette tapes require equalization of 3180 p5 (50
Hz) and 120 ps (1.3 kHz). These time constants result in an
AC gain at 1 kHz given by:
AV=25[
- R4R3
Av (1 kHz) = 25 (a-sir-it-lt-il-a,-
) 1 .663 (2)
3180 us or 50 Hz
120 'M' or 1326 Hz
Using the pole and zero locations of the transfer function,
the two other equations needed to solve for the component
values are:
4 = 2erC1(1326 Hz) (3)
- 2erC1(50 Hz) 21:01 (1326 Hz) - 2erC1 (51.96)
R3 (4)
We can now solve for C1 as a function of H2, or:
[27TC1(1326)] [2701(51.96)]
Avil kHz) = -25 (1.663)
[R2 21rC1(50)] (5)
= -4.80 M 10-3 6
R2 [Av (1 kHz)] ( )
When chromium dioxide is used, the defined time constants
are 3180 p9 and 70 11.5. This changes equation (3) to:
4 - 2rC1(2274 Hz) (7)
The value of R3 is normally not changed. This results in an
error of less than 0.2 dB in the low frequency response.
The output voltage of the LM1837 is set by the input amplifi-
er DC voltage at pin 5 or 14, and by R3 and R5.
Pins 1 and 18 are biased 0.7V less than VouT(pin 2 or 17).
When these diodes are used the output (pin 2 or 17) should
be biased at one half the minimum operating supply voltage.
Equation (8) can be rewritten to solve for R5.
2.5 R3
R5 Vo - 2.5 (9)
The output voltage of the LM1837 will vary from that given
in equation (8) due to variations in the input amplifier DC
voltage as well as the output amplifier input bias current,
input offset current and input offset voltage. The following
equation gives the worst-case variation in the output voltage
in either forward or reverse state.
AVOUT = i [AVPIN3 (1 + 'e) +
Nominal VOUT (pin 2 or 17) = 2.5 (1 + E) (8)
tra (AIBIA5(R1 - R2) + $611 + R2) + vos)]
Using the worst-case values in the electrical characteristics
reduces this to
AVOUT = l [0.4 (1 + Fl) +
R5 (11)
E (200 nA(Ft1 - R2) + 50nA(Ft1+ R2) + 5 mv)]
Equation (10) does not incorporate the effect of mute resis-
tors on the output voltage. The presence of mute resistors
causes an additional offset
AV(pins 5-14)
AVOUT(mute) 2(R1 + RT) R1 (12)
For the circuit in Figure t worst-case:
AVouT(mute) = m A 1.5M = IV
2(20k + 270k)
This means that the output pins 2 and 17 would differ by IV.
The trade off here is the amount of unmute voltage versus
the DC accuracy of pins 2 and 17.
LM1837
Application Hints (Continued)
The turn-on delay is set by R1 and C2; delay can be approx-
imated by:
eaytmet R1C2 Ln (V )(R ) (13)
EXAMPLE
If we desire a tape preamp with 100 mV output signal from a
tape head with a nominal output of 0.5 mV at 1 kHz for
standard ferric cassette tape, the external components are
determined as follows. The value of R2 is arbitrarily set to
10 kn.
R1 =R2= 10k
This minimizes errors due to the output amplifier bias cur-
rents.
-4.80 M 10-3
- 100 mv]
0.5 mV
Use 0.0022 “F and determine:
CI = = 2400 pF - 0.0022 WF
1tsknl
R4 = = 54.6 kn - 54.9 kft 1%
2ertM(1326)
3 = -....1.,
thrC1(51.96)
To bias the output amplifier output voltage at 6V (half sup-
--.. 1.39 Mn-r 1.4Mn1%
_ 2.5(1.4 Mn)
6 - 2.5
The maximum variation in the output is found using equation
R5 =1MQ
AVOUT = 11.9V
The low frequency response and turn-on delay detemine
the value of C2. For R1 = 10k and C2 = 10 MP the low
frequency 3 dB point is 1.6 Hz and the turn-on delay is 0.4
seconds, from equation (12).
The complete circuit is shown in Figure P, A circuit with 5%
components and biased for a minimum supply of 10V is
shown in Figure t. If additional gain is needed R1 and R2
can be reduced without changing the frequency response of
the circuit.
DIODE SWITCHING
The LM1837 has a diode in series with each output for
source switching applications. The outputs of several func-
tional blocks can be diode OR-connected as shown in Fig-
ure 9.
By removing the power supply from the FM demodulator, its
output diode will be cut off by the LM1837 output DC volt.
age. R6 is used to bias ON the diode of the LM1837 when
power is applied to it. When the output is taken from pin 1 or
pin 18, the THD will be higher because of the current modu-
lation in the diode.
TL/H/7902-18
FIGURE 9
CROSSTALK AND CHANNEL SEPARATION
When two signal sources share a common reference point
which is separated from ground by a resistance, there will
always be some amount of interchannel crosstalk (the re-
ciprocal of channel separation) induced. The coupling meth-
od of Figure f is examined to determine whether the in-
duced crosstalk is acceptably low.
Figure " is the equivalent AC circuit for the connection
scheme of Figure f. RB is the Thevenin resistance of the
common bias point, R[N is the preamplifier input resistance,
2s is the impedance of the playback head, and V57. V38.
V311. and V312 are the open-circuit output voltages of the
sources. If we set vss V311, and V512 equal to zero, we
can define crosstalk for this circuit as V12/V7, where V7
and V12 are the AC signal voltages appearing at the two
preamplifierinputs, assuming RB < RIN/S.
The crosstalk can be shown to be:
I/I., - Re -
V7 RB + 28 + RIN/a
Since its is dependent on the measurement frequency and
the particular head used, we choose the worst-case condi-
tion and set its = O. The minimum value of RN is 150 kn,
and RB =1 1000. This yields a crosstalk figure ot:
V12 100
V7 _ 50100 - -54dB
This is 14 dB better than the minimum guaranteed channel
separation, so the connection method of Figure 1 will pro-
vide acceptable crosstalk levels.
Reference 1: CClR/ARM: A Practical Noise Measurement
Method; by Ray Dolby, David Robinson and Kenneth Gun-
dry, AES Preprint No, 1353 (F-3).
Application Hints (Continued)
FORMRD
FIGURE 10. Att Equivalent of Figure t
TL/H/7902-19
1.88”“
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This file is the datasheet for the following electronic components:
LM1837N - product/lm1837n?HQS=TI-nu|I-nu|I-dscatalog-df-pf-null-wwe
LM1837M - product/lm1837m?HQS=T|-nu|I-nu|I-dscataIog-df-pf-null-wwe
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